Comparison circuit and analog-to-digital conversion device

ABSTRACT

A comparison circuit includes: an input circuit includes a first transistor for receiving a first signal, and a second transistor for receiving a second signal; a first current route of which the electric current is controlled by the first transistor; a second current route of which the electric current is controlled by the second transistor; a latch for amplifying potential difference between the first current route and the second current route; a comparative operation control circuit including a first switch for executing or blocking supply voltage to the drain of the first transistor, a second switch for executing or blocking supply voltage to the drain of the second transistor, and a third switch for executing supply voltage to the first current route and the second current route; a comparative operation setting circuit for controlling supply or blocking of supply of the first switch, the second switch, and the third switch.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is based upon and claims the benefit of priority fromthe prior Japanese Patent Application NO. 2009-074286 filed on Mar. 25,2009, and the prior Japanese Patent Application NO. 2009-149496 filed onJun. 24, 2009, the entire contents of which are incorporated herein byreference.

FIELD

The embodiments discussed herein are related to a comparison circuit, ananalog-to-digital conversion device using the comparison circuitthereof, and a signal processing device using the analog-to-digitalconversion device thereof.

BACKGROUND

Voltage-comparative-type comparators frequently used for a controlcircuit and the like generally include two MOS transistors of which thegates receive one of differential input signals, two current routes ofwhich the currents are controlled by the MOS transistors thereofaccording to the voltage of the differential input signal, and a latchunit configured to amplify and hold potential difference between thecurrent routes.

Accordingly, in the event of executing comparison between the voltagesof the differential input signals with the above comparator according tothe difference of properties of the above MOS transistors, or theamplification and holding property of the latch unit, error occurs. As aresult thereof, conversion error occurs in an analog-to-digitalconverter configured of this voltage-comparative-type comparator.

Related art is discussed in P. M. Figueiredo, P. Cardoso, A. Lopes, C.Fachada, N. Hamanishi, K. Tanabe, and J. Vital, “A 90 nm CMOS 1.2V 6b1GS/s Two-Step Subranging ADC,” IEEE International Solid-State CircuitsConference, Session 31/31.2, February 2006, and J. Craninckx and G. Vander Plas, “A 65 fJ/Conversion-Step 0-to-50MS/s 0-to-0.7 mW 9bCharge-Sharing SAR ADC in 90 nm Digital CMOS,” IEEE InternationalSolid-State Circuits Conference, Session 13/13.5, vol. XL, pp. 246-247,600, February 2007.

SUMMARY

According to one aspect of the embodiments includes: an input circuitmade up of a first transistor configured to receive a first signal at agate electrode, and a second transistor configured to receive a secondsignal at a gate electrode; a first current route of which the currentis controlled by the first transistor according to the voltage of thefirst signal; a second current route of which the current is controlledby the second transistor according to the voltage of the second signal;a latch circuit configured to amplify potential difference between afirst node within the first current route and a second node within thesecond current route; a comparative operation control circuit includinga first switch configured to execute supply of high potential or supplyof ground potential to the drain of the first transistor, or supply ofhigh potential or blocking of supply of ground potential to the drain, asecond switch configured to execute supply of high potential or supplyof ground potential to the drain of the second transistor, or supply ofhigh potential or blocking of supply of ground potential to the drain,and a third switch configured to execute supply or blocking of supply ofground potential to the first current route and the second currentroute; and a comparative operation setting circuit configured toindependently control the supply or blocking of supply of the firstswitch, the second switch, and the third switch.

The object and advantages of the embodiments will be realized andattained by means of the elements and combinations particularly pointedout in the claims.

It is to be understood that both the foregoing general description andthe following detailed description and are exemplary and explanatory andare not restrictive of the embodiments, as claimed.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a circuit diagram illustrating a comparison circuit accordingto a first embodiment.

FIGS. 2A, 2B, 2C, and 2D are circuit diagrams illustrating a delaycircuit and a logic circuit.

FIG. 3 illustrates a flowchart relating to the operation of the logiccircuit for controlling the delay circuit.

FIG. 4 illustrates a timing chart for describing the operation of thelock circuit.

FIGS. 5A and 5B are diagrams representing change in the signalpotentials of signals when delaying the leading-edge point-of-time of asignal.

FIG. 6 is a circuit diagram illustrating a comparison circuit accordingto a second embodiment.

FIG. 7 is a circuit diagram illustrating a comparison circuit accordingto a third embodiment.

FIG. 8 is a circuit diagram illustrating a comparison circuit accordingto a fourth embodiment.

FIG. 9 is a circuit diagram illustrating a comparison circuit accordingto a fifth embodiment.

FIG. 10 illustrates an ADC (Analog Digital Converter) according to asixth embodiment.

FIGS. 11A and 11B illustrate examples of inverters of a delay circuit.

FIGS. 12A and 12B are diagrams for describing operation according to thesixth embodiment regarding a latch unit, an input unit, a comparativeoperation control circuit, and the delay circuit.

FIG. 13 is a diagram representing relationship of the logic of a signal,trailing time difference of signals, and potential difference betweensignals making up an input signal.

FIG. 14 illustrates operation waves when detecting potential differenceof complementary signals made up of signals according to the ADC circuitaccording to the sixth embodiment.

FIG. 15 is a flowchart for describing the control of a calculationexecuted by a successive comparative operation control circuit, and adetection method of difference input potentials executed by the controlthereof.

FIGS. 16A and 16B are tables for describing a method for derivingrelationship between a binary numeral represented by a signal, and abinary numeral represented by a digital signal to be output byanalog-to-digital conversion by the ADC in the case that there is nolinearity with correlation as to difference between the potentials ofsignals.

FIG. 17 illustrates an ADC according to a seventh embodiment.

FIG. 18 illustrates an ADC circuit according to an eighth embodiment.

FIG. 19 illustrates an ADC circuit according to a ninth embodiment.

FIG. 20 is a diagram illustrating an ADC according to a tenthembodiment.

FIG. 21 is a diagram illustrating a reception device using the ADCsillustrated by the sixth through tenth embodiments.

DESCRIPTION OF EMBODIMENTS

The present invention encompasses modifications, obtained by addingdesign modifications to the embodiments described below, which oneskilled in the art can conceive, and modifications obtained by executingrecombination of components in the embodiments. Also, the presentinvention also encompasses modifications wherein the components arereplaced with other components which yield the same operations, effects,or the like, and the present invention is not restricted to thefollowing embodiments.

First Embodiment

FIG. 1 is a circuit diagram illustrating a comparison circuit 10according to the first embodiment. The comparison circuit 10 includes alatch unit 20, an input unit 30, a comparative operation control circuit40, and a comparative operation setting circuit 50.

The comparative operation setting circuit 40 includes P-type MOStransistors 42 and 43, and an N-type MOS transistor 41.

The N-type MOS transistor 41 has a drain to be connected to the sourcesof N-type MOS transistors 31 and 32 of the input unit 30, a source to beconnected to ground VSS 70, and a gate for receiving a signal L. TheN-type MOS transistor 41 supplies the ground potential from the groundVSS 70 to the input unit 30 when the logic of the signal L is “H”, andblocks supply of ground potential form the ground VSS 70 to the inputunit 30 when the logic of the signal L is “L”. The N-type MOS transistor41 serves as a switch for connecting or blocking the input unit 30 andthe ground VSS 70.

The P-type MOS transistor 42 has a source to be connected to ahigh-potential VDD power source 60, a drain to be connected to thesource of an N-type MOS transistor 23 of the latch unit, and a gate forreceiving a signal LM.

The P-type MOS transistor 42 blocks supply of high potential VDD fromthe high-potential VDD power source 60 to the latch unit 20 and theinput unit 30 when the logic of the signal LM is “H”, and supplies thehigh-potential VDD from the high-potential VDD power source 60 to thelatch unit 20 and the input unit 30 when the logic of the signal LM is“L”. The P-type MOS transistor 42 serves as a switch for connecting orblocking the latch unit 20 and the input unit 30, and the high-potentialVDD power source 60.

The P-type MOS transistor 43 has a source to be connected to thehigh-potential VDD power source 60, a drain to be connected to thesource of an N-type MOS transistor 24 of the latch unit, and a gate forreceiving a signal LP.

The P-type MOS transistor 43 blocks supply of high potential VDD fromthe high-potential VDD power source 60 to the latch unit 20 and theinput unit 30 when the logic of the signal LP is “H”, and supplies thehigh-potential VDD from the high-potential VDD power source 60 to thelatch unit 20 and the input unit 30 when the logic of the signal LM is“L”. The P-type MOS transistor 43 serves as a switch for connecting orblocking the latch unit 20 and the input unit 30, and the high-potentialVDD power source 60.

The input unit 30 includes N-type MOS transistors 31 and 32. The N-typeMOS transistor 31 has a drain to be connected to the source of theN-type MOS transistor 23 of the latch unit, a source to be connected tothe drain of the N-type MOS transistor 41 of the comparative operationcontrol circuit 40, and a gate for receiving an input signal VIP. The onresistance value of the N-type MOS transistor 31 varies according to thepotential of the input signal VIP. The N-type MOS transistor 32 has adrain to be connected to the source of an N-type MOS transistor 24 ofthe latch unit, a source to be connected to the drain of the N-type MOStransistor 41 of the comparative operation control circuit 40, and agate for receiving a signal VIM. The on resistance value of the N-typeMOS transistor 32 varies according to the potential of the signal VIM.

As a result thereof, signals obtained by inverting the logics of theinput signals VIP and VIM occur on the drains of the N-type MOStransistors 31 and 32, respectively.

The latch unit 20 includes P-type MOS transistors 21 and 22, and theN-type MOS transistors 23 and 24.

The P-type MOS transistor 21 has a drain to be connected to the drain ofthe N-type MOS transistor 23, a gate to be connected to the drain of theP-type MOS transistor 22, and a source to be connected to thehigh-potential VDD power source 60.

The P-type MOS transistor 22 has a drain to be connected to the drain ofthe N-type MOS transistor 24, a gate to be connected to the drain of theP-type MOS transistor 23, and a source to be connected to thehigh-potential VDD power source 60.

The N-type MOS transistor 23 has a drain to be connected to the drain ofthe P-type MOS transistor 21, a gate to be connected to the drain of theN-type MOS transistor 24, and a source to be connected to the drain ofthe N-type MOS transistor 31 of the input unit 30. As a result thereof,a signal obtained by inverting the logic of the input signal VIP appearson the drain of the N-type MOS transistor 23.

The N-type MOS transistor 24 has a drain to be connected to the drain ofthe P-type MOS transistor 22, a gate to be connected to the drain of theN-type MOS transistor 23, and a source to be connected to the drain ofthe N-type MOS transistor 32 of the input unit 30. As a result thereof,a signal obtained by inverting the logic of the input signal VIM appearson the drain of the N-type MOS transistor 24.

Output signals OM and OP are output from the latch unit 20. The outputsignal OM is connected to a node A between the drain of the P-type MOStransistor 21 and the drain of the N-type MOS transistor 23. The outputsignal OP is connected to a node B between the drain of the P-type MOStransistor 22 and the drain of the N-type MOS transistor 24.

The gate of the P-type MOS transistor 21 and the drain of the P-type MOStransistor 22 of the latch unit 20 are connected to the node B, and thegate of the P-type MOS transistor 22 and the drain of the P-type MOStransistor 21 are connected to the node A. That is to say, the P-typeMOS transistor 21 and the P-type MOS transistor 22 are connectedcrosswise to the nodes A and B, and accordingly, the P-type MOStransistor 21 and the P-type MOS transistor 22 amplify potentialdifference between the node A and node B.

The gate of the N-type MOS transistor 23 and the drain of the N-type MOStransistor 24 of the latch unit 20 are connected to the node B, and thegate of the N-type MOS transistor 24 and the drain of the N-type MOStransistor 23 are connected to the node A. That is to say, the N-typeMOS transistor 23 and the N-type MOS transistor 24 are connectedcrosswise to the nodes A and B, and accordingly, the N-type MOStransistor 23 and the N-type MOS transistor 24 amplify potentialdifference between the node A and node B.

Thus, the P-type MOS transistor 21 and N-type MOS transistor 23 of thelatch unit 20, and N-type MOS transistor 31 are connected seriallybetween the high-potential VDD power source 60 and the drain of theN-type MOS transistor 41, and make up a first current route includingthe node A. The P-type MOS transistor 22 and N-type MOS transistor 24 ofthe latch unit 20, and N-type MOS transistor 32 are connected seriallybetween the high-potential VDD power source 60 and the drain of theN-type MOS transistor 41, and make up a second current route includingthe node B.

Therefore, when the signals L, LM, and LP are “H”, supply of thehigh-potential VDD to the latch unit 20 and the input unit 30 is blockedby the P-type MOS transistors 42 and 43, and ground potential issupplied to the input unit 30 by the N-type MOD transistor 41. In theabove case, the potentials of the nodes A and B start to decrease fromthe high-potential VDD. The on resistance of the N-type MOS transistor31 varies according to the potential of the input signal VIP, and the onresistance of the N-type MOS transistor 32 varies according to thepotential of the input signal VIM, and accordingly, the current amountflowing to each current route varies. Thus, the rates of decrease at thenodes A and B vary. Of the potentials of the nodes A and B, one firstreaching the threshold of the latch unit 20 becomes “L”. If we say thatthe potential of the node A first reaches the threshold of the latchunit 20, the P-type MOS transistor 22 is turned on, and the potential ofthe node B increases, and the logic thereof becomes “H”. Conversely, theP-type MOS transistor 21 is turned off, and the potential of the node Adecreases, and the logic thereof becomes “L”.

Note that, when the signals L, LM, and LP are “L”, the high-potentialVDD is supplied to the latch unit 20 and the input unit 30 by the P-typeMOS transistors 42 and 43, and supply of the ground potential to theinput unit 20 is blocked by the N-type MOS transistor 41.

As a result thereof, the potential difference between the nodes A and Bis 0, or almost eliminated.

The comparative operation setting circuit 50 includes a delay circuit51, and a logic circuit 56 for controlling the delay circuit 51.

FIGS. 2A, 2B, 2C, and 2D are circuit diagrams illustrating the delaycircuit 51 and the logic circuit 56.

FIG. 2A is a circuit diagram illustrating the delay circuit 51. Thedelay circuit 51 includes a delay circuit 52, a delay circuit 53, andAND circuits 54 and 55.

The delay circuit 52 receives the signal L, and outputs a signal LPAillustrating change in potential wherein the delay corresponding to abinary numeral DCP represented by digital signals DCP<0> through DCP<2>is added to change in the potential of the signal L. Note that thebinary numeral DCP represented by the digital signals DCP<0> throughDCP<2> advances, such as shown later, from (111) to (000), and whenchanging from the digital signal DCM<0> to the digital signal DCM<2>,maintains (000).

The delay circuit 53 receives the signal L, and outputs a signal LMAillustrating change in potential wherein the delay corresponding to thebinary numeral DCM represented by the digital signals DCM<0> throughDCM<2> is added to change in the potential of the signal L. Note thatthe binary numeral DCM represented by the digital signals DCM<0> throughDCM<2> advances, such as shown later, from (000) to (111), and whenchanging from the digital signal DCM<0> to the digital signal DCM<2>,maintains (111).

The AND circuit 54 outputs a signal LP having the logical AND betweenthe logic of a signal LPA and the logic of the signal L. The AND circuit55 outputs a signal LM having the logical AND between the logic of asignal LMA and the logic of the signal L. Accordingly, the signal LP ischanged from the logic “L” to logic “H” generally at the same time asthe signal LPA being changed from the logic “L” to logic “H”. On theother hand, the signal LP is changed from the logic “H” to logic “L”generally at the same time as the signal L being changed from the logic“H” to logic “L”. The signal LM is changed from the logic “L” to logic“H” generally at the same time as the signal LMA being changed from thelogic “L” to logic “H”. On the other hand, the signal LM is changed fromthe logic “H” to logic “L” generally at the same time as the signal Lbeing changed from the logic “H” to logic “L”.

Thus, upon a switch made up of the N-type MOS transistor 41 being turnedon by the signal L, a switch made up of the P-type MOS transistor 42 isturned on with delay according to the binary numeral DCM represented bythe digital signals DCM<0> through DCM<2>, a switch made up of theP-type MOS transistor 43 is turned on with delay according to the binarynumeral DCP represented by the digital signals DCP<0> through DCP<2>.

FIG. 2B is a circuit diagram illustrating a first example 58 of thedelay circuits 52 and 53. The first example 58 of the delay circuits 52and 53 includes a P-type MOS transistor 580 and an N-type MOS transistor581 which make up an inverter 58A, a P-type MOS transistor 588 and anN-type MOS transistor 589 which make up an inverter 58B, inverters 585,586, and 587, and capacitances 582, 583, and 584 made up of a gateelectrode-insulating film-substrate electrode-type transistor, i.e., aso-called MOS-type transistor. The inverter 58A outputs a signalobtained by inverting the logic of the signal L to the inverter 58B. Theinverter 58B receives the inverted signal of the signal L, and outputsthe signal LPA having further inverted logic thereof (the signal LPA atthe time of the first example 58 corresponding to the delay circuit 52,and the signal LMA when the first example corresponding to the delaycircuit 53). The capacitances 582, 583, and 584 are connected to asignal line which connects the inverter 58A and the inverter 58B by thegate electrode. When the potential of the substrate electrode is high,and the threshold voltage of the MOS-type transistor is equal to orgreater than the ground voltage, the capacitances 582, 583, and 584 havegreat capacitance according to the thickness of the insulating film, butwhen the potential of the substrate electrode is low, and the thresholdvoltage of the MOS-type transistor is less than the ground voltage, havesmall capacitance. In the event that the electrostatic capacitances ofthe capacitances 582, 583, and 584 are compared with those at the timeof having the same threshold voltage, if we say that the capacitance 582is 1, the capacitance 583 is the ratio of 2, and the capacitance 584 isthe ratio of 4. The outputs of the inverters 585, 586, and 587 areconnected to the substrate electrodes of the capacitances 582, 583, and584, respectively. The inputs of the inverters 585, 586, and 587 areconnected to the digital DCP<0> through DCP<2>, respectively (note that,when the first example 58 corresponds to the delay circuit 53, these areconnected to the DCM<0> through DCM<2>). Accordingly, the signal fromthe inverter 58A to the inverter 58B is delayed according to the binarynumeral represented by the digital signals DCP<0> through DCP<2>.

FIG. 2C is a circuit diagram illustrating a second example 59 of thedelay circuits 52 and 53. The second example 59 of the delay circuits 52and 53 includes a P-type MOS transistor 590 and an N-type MOS transistor591 which make up an inverter 59A, a P-type MOS transistor 592 and anN-type MOS transistor 593 which make up an inverter 59B, a variableresistor 594 connected to the P-type MOS transistor 590 and thehigh-potential VDD power source 60 and serially connected to the P-typeMOS transistor 590, and a variable resistor 595 connected to the N-typeMOS transistor 591 and the ground power source and serially connected tothe N-type MOS transistor 591. The inverter 59A receives the signal L,and outputs an inverted signal thereof to the inverter 59B. The inverter59B further inverts the signal from the inverter 59A to take this as thesignal LPA (the signal LMA when the second example 59 represents thedelay circuit 53). With the variable resistors 594 and 595, theresistance varies according to the binary numeral DCP represented by thedigital signals DCP<0> through DCP<2>, and become the maximum resistancewhen the binary numeral DCP is (111) (when the second example 59represents the delay circuit 53, the resistance varies according to thebinary numeral DCM represented by the digital signals DCM<0> throughDCM<2>). Accordingly, the potential supplied to the inverter 59Aincreases or decreases according to the variable resistors 594 and 595,and accordingly, the signal to be output from the inverter 59A is outputwith delay according the potential thereof from the input signal.

FIG. 2D is a circuit diagram illustrating a specific example of thelogic circuit 56 for controlling the delay circuit 51. The logic circuit56 includes a DFF 560, DFF 570, JKFF 561, JKFF 562, JKFF 563, JKFF 564,JKFF 571, JKFF 572, JKFF 573, JKFF 574, AND 565, 566, 567, 568, 569,575, 576, 577, and 579.

Note that JKFF means a JK flip-flop. Specifically, when J=“H” and K=“H”,signals to be output from terminals Q and /Q are logically inverted eachtime clock enters a clock terminal CK. Also, when J=“L” and K=“L”, thelogics of the signals to be output from the terminals Q and /Q areunchanged. Further, when J=“L” and K=“H”, the terminal Q outputs “H”,and the terminal /Q outputs “L”, and when J=“H” and K=“L”, the terminalQ outputs “L”, and the terminal /Q outputs “H”. Note that according tothe definitions of the terminals /Q and Q, the logics of the outputsignals of the terminals Q and /Q when J=“L” and K=“H”, and the logicsof the output signals of the terminals Q and /Q when J=“H” and K=“L”counterchange. In this case, let us say that the signals DCP<0> throughDCP<2>, and DCM<0> through DCM<2> counterchange.

The logic circuit 56 receives signals CS, CD, CK, and OP, and outputsthe signals DCP<0> through DCP<2>, and DCM<0> through DCM<2>.

The AND 568 receives input of the signals CS and OP, and outputs asignal having a logic obtained by logical AND of both signals toterminals D of the DFF 560 and 570. Accordingly, when the signal CSrises from logic “L” to logic “H”, the signal OP has originally logic“H”, the AND 568 outputs the signal of logic “H”. Subsequently, when thesignal OP becomes logic “L”, the AND 568 outputs the signal of logic“L”.

The DFF 560 receives a clock signal CK from the terminal CK, latches thelogic of the input signal from the terminal D at the leading edge of theclock signal CK, and outputs the latched signal from the terminal Q.

The AND 569 receives the signal from the terminal Q of the DFF 560, andthe signal /CE, and outputs a signal having a logic obtained by logicalAND of both signals to the JKFF 561, JKFF 562, JKFF 563, and JKFF 564.Here, the signal /CE is a signal having the inverted logic of the signalCE. Accordingly, when the signal from the terminal Q of the DFF 560 islogic “H”, the output signal of the AND 569 has the same logic as thelogic of the signal /CE. On the other hand, when the signal from theterminal Q of the DFF 560 is logic “L”, the signal of logic “L” isoutput to terminals k of the JKFF 561, JKFF 562, JKFF 563, and JKFF 564.

The JKFF 561 receives the latched signal from the terminal Q of the DFF560 by a terminal J, receives the output of the AND 569 by the terminalk, and receives the clock signal CK by the terminal CK. The JKFF 562receives the latched signal from the terminal Q of the DFF 560 by theterminal J, receives the output of the AND 569 by the terminal k, andreceives the output signal from the terminal Q of the JKFF 561 by theterminal CK. The JKFF 563 receives the latched signal from the terminalQ of the DFF 560 by the terminal J, receives the output of the AND 569by the terminal k, and receives the output signal from the terminal Q ofthe JKFF 562 by the terminal CK. The JKFF 564 receives the latchedsignal from the terminal Q of the DFF 560 by the terminal J, receivesthe output of the AND 569 by the terminal k, and receives the outputsignal from the terminal Q of the JKFF 563 by the terminal CK. The AND565 receives the signal from the terminal Q of the JKFF 561, and thesignal from the terminal Q of the JKFF 564, and output a signal obtainedby logical AND thereof as the digital signal DCP<0>. The AND 566receives the signal from the terminal Q of the JKFF 562, and the signalfrom the terminal Q of the JKFF 564, and output a signal obtained bylogical AND thereof as the digital signal DCP<1>. The AND 567 receivesthe signal from the terminal Q of the JKFF 563, and the signal from theterminal Q of the JKFF 564, and output a signal obtained by logical ANDthereof as the digital signal DCP<2>.

The DFF 570 receives the clock signal CK from the terminal CK, latchesthe logic of the input signal from the terminal D at the leading edge ofthe clock signal CK, and outputs the latched signal from the terminal Q.

The AND 579 receives the signal from the terminal Q of the DFF 570, andthe signal /CE, and outputs a signal having a logic obtained by logicalAND of both signals to the JKFF 571, JKFF 572, JKFF 573, and JKFF 574.Accordingly, when the signal from the terminal Q of the DFF 570 is logic“H”, the output signal of the AND 579 has the same logic value as thelogic of the signal /CE. On the other hand, when the signal from theterminal Q of the DFF 570 is logic “H”, the signal of logic “L” isoutput to the terminals k of the JKFF 571, JKFF 572, JKFF 573, and JKFF574.

The JKFF 571 receives the latched signal from the terminal Q of the DFF570 by the terminal J, receives the output of the AND 579 by theterminal k, and receives the clock signal CK by the terminal CK. TheJKFF 572 receives the latched signal from the terminal Q of the DFF 570by the terminal J, receives the output of the AND 579 by the terminal k,and receives the output signal from the terminal Q of the JKFF 571 bythe terminal CK. The JKFF 573 receives the latched signal from theterminal Q of the DFF 570 by the terminal J, receives the output of theAND 579 by the terminal k, and receives the output signal from theterminal Q of the JKFF 572 by the terminal CK. The JKFF 574 receives thelatched signal from the terminal Q of the DFF 570 by the terminal J,receives the output of the AND 579 by the terminal k, and receives theoutput signal from the terminal Q of the JKFF 573 by the terminal CK.The AND 575 receives the signal from the terminal Q of the JKFF 571, andthe signal from the terminal Q of the JKFF 574, and outputs a signalobtained by logical AND thereof as the digital signal DCM<0>. The AND576 receives the signal from the terminal Q of the JKFF 572, and thesignal from the terminal Q of the JKFF 574, and outputs a signalobtained by logical AND thereof as the digital signal DCM<1>. The AND577 receives the signal from the terminal Q of the JKFF 573, and thesignal from the terminal Q of the JKFF 574, and outputs a signalobtained by logical AND thereof as the digital signal DCM<2>.

Thus, when the signal CS and the signal OP are logic “H”, the AND 568outputs the signal of logic “H” to the terminal D of the DFF 560. Whenreceiving logic “H” at the terminal D, upon the clock signal CK beinginput, the DFF 560 outputs the signal of logic “H” by the terminal Q. Asa result thereof, the combination of digital signals to be output fromthe terminal Q of each of the JKFF 561, 562, and 563 is counted downfrom (1, 1, 1) to (0, 0, 0). While the combination of the above digitalsignals is counted down from (1, 1, 1) to (0, 0, 0), the signal of logic“H” is output from the terminal Q of the JKFF 564, and accordingly, thesignal logic of “H” is input to one of the inputs of the AND 565, 566,and 567, and accordingly, the combination from the digital signalsDCP<0> through DCP<2> is counted from (1, 1, 1) to (0, 0, 0). Upon thecountdown ending and reaching (0, 0, 0), the signal of logic “L” isoutput from the terminal Q of the JKFF 564, and accordingly, regardlessof the logic of the signal output from the terminal Q of each of theJKFF 561, 562, and 563, the combination from the digital signals DCP<0>to DCP<2> is also held in (0, 0, 0).

Note that during the above countdown upon the logic of the signal OPbecoming logic “L”, the logic of the signal from the terminal Q of theDFF 570 becomes logic “L”, and accordingly, the countdown of thecombination from the digital signals DCP<0> to DCP<2> is ended, and thevalues thereof are maintained.

On the other hand, the combination of the digital signals output fromthe terminal of each of the JKFF 571, 572, and 573 is counted up from(0, 0, 0) to (1, 1, 1). However, while the combination from the digitalsignals DCP<0> to DCP<2> is counted down from (1, 1, 1) toward (0, 0,0), the signal of logic “L” is output from the terminal Q of the JKFF574, and accordingly, the signal logic “L” is input to one of the inputsof the AND 575, 576, and 577, the combination from the digital signalsDCP<0> to DCP<2> is held in (0, 0, 0). However, upon the countdownending, and the combination from the digital signals DCP<0> to DCP<2>reaching (0, 0, 0), the signal of logic “H” is output from the terminalQ of the JKFF 574, and accordingly, the combination from the digitalsignals DCM<0> to DCM<2> is also counted from (0, 0, 0) toward (1, 1, 1)according to the logic of the signal to be output from the terminal Q ofeach of the JKFF 571, 572, and 573.

Note that during the above count-up upon the logic of the signal OPbecoming logic “L”, the logic of the terminal Q of the DFF 570 becomeslogic “L”, and accordingly, the count-up of the combination from thedigital signals DCM<0> to DCM<2> ends, and the values thereof aremaintained.

FIG. 3 illustrates a flowchart relating to the operation of the logiccircuit 56 for controlling the delay circuit 51. A specific example ofthe logic circuit 56 is illustrated in FIG. 2D, but it goes withoutsaying that as long as a circuit which operates in accordance with theflowchart in FIG. 3, this circuit may be configured in any wise.

With operation OP1, determination is made whether or not the logic ofthe signal CS rises to “1” (logic “H”). When the logic of the signal CSrises to “1”, the logic circuit 56 starts its operation, andaccordingly, executes the next operation OP2. When the logic of thesignal CS is “0” (logic “L”), the logic circuit 56 maintains an idlestate.

With operation OP2, upon the logic of the signal CE being set to “0”,i.e., the logic of the signal /CE being set to “1”, so that the binarynumeral DCP represented by the digital signals DCP<0> to DCP<2> becoming(111), the logic circuit 56 outputs these digital signals. Also, so thatthe binary numeral DCM represented by the digital signals DCM<0> toDCM<2> becoming (000), the logic circuit 56 outputs these digitalsignals.

With operation OP3, after the logic of the signal CS rises to “1”,determination is made whether or not the logic of the clock signal CKrises to “1”. In the case that the logic of the clock signal CK is “1”,the logic circuit 56 maintains its state.

With operation OP4, the logic circuit 56 determines whether or not theclock signal CK rises from “0” to “1”. In the case that the clock signalCK does not rise from “0” to “1”, the logic circuit 56 maintains itsstate. In the case that the clock signal CK rises from “0” to “1”, thelogic circuit 56 proceeds to the next operation OP5.

With operation OP5, the logic circuit 56 determines whether or not thelogic of the signal OP is “1”. In the case that the logic of the signalOP is “0”, the logic circuit 56 externally outputs a signal indicatingthat the logic of the signal OP is “0”, and receives the signal CE ofwhich the logic is “1”. In the case that the logic of the signal OP is“1”, the logic circuit 56 proceeds to the next operation OP6.

With operation OP6, the logic circuit 56 determines whether or not thebinary numeral DCP represented by the digital signals DCP<0> to DCP<2>is (000). When the binary numeral DCP represented by the digital signalsDCP<0> to DCP<2> is (000), the logic circuit 56 proceeds to the nextoperation OP8. When the binary numeral DCP represented by the digitalsignals DCP<0> to DCP<2> is not (000), the logic circuit 56 proceeds tothe next operation OP7.

With operation OP7, the logic circuit 56 executes operation wherein 1 issubtracted from the binary numeral DCP represented by the digitalsignals DCP<0> to DCP<2>. That is to say, the logic circuit 56 executesthe countdown operation of the binary numeral represented by the digitalsignals DCP<0> to DCP<2>. Subsequently, the logic circuit 56 proceeds tooperation OP4.

With operation OP8, the logic circuit 56 adds 1 to the binary numeralDCM represented by the digital signals DCM<0> to DCM<2>, i.e., counts upthis binary numeral.

With operation OP9, the logic circuit 56 determines whether or not theclock signal CK rises from “0” to “1”. In the case that the clock signalCK does not rise from “0” to “1”, the logic circuit 56 maintains itsstate. In the case that the clock signal CK rises from “0” to “1”, thelogic circuit 56 proceeds to the next operation OP10.

With operation OP10, the logic circuit 56 determines whether or not thelogic of the signal OP is “1”. In the case that the logic of the signalOP is “0”, the logic circuit 56 externally outputs a signal indicatingthat the logic of the signal OP is “0”, and receives the signal CE ofwhich the logic is “1”. In the case that the logic of the signal OP is“1”, the logic circuit 56 proceeds to the next operation OP11.

With operation OP11, the logic circuit 56 determines whether or not thebinary numeral DCM represented by the digital signals DCM<0> to DCM<2>is (111). In the case that the binary numeral DCM is (111), the logiccircuit 56 externally outputs a signal indicating that the logic of thesignal OP is “0”, and receives the signal CE of which the logic is “1”.In the case that the binary DCM is not (111), the logic circuit 56proceeds to operation OP12.

With operation OP12, the logic circuit 56 adds 1 to the binary numeralDCM represented by the digital signals DCM<0> to DCM<2>, i.e., counts upthis binary numeral.

FIG. 4 illustrates a timing chart for describing the operation of thelogic circuit 56.

The logic of the signal CS changes from logic “L” to logic “H” atpoint-in-time T0, and maintains logic “H” even at point-in-time Tm+2.

The signal CE changes from logic “H” to logic “L” at point-in-time T0,and changes from logic “L” to logic “H” at point-in-time Tm+2.

(VIP-VIM) indicates the voltage difference between the signals VIP andVIM, (VIP-VIM) is 0 while the logic of the signal CS is “L”.

The clock signal CK is a signal which repeats the periods of logic “H”and logic “L”, and the periods of logic “H” and logic “L” have generallythe same length.

The signal L is synchronized with the reversed phase of the clock signalCK, and also the periods of logic “H” and logic “L” are generally thesame as with the clock signal CK.

The binary numeral DCP is made up of the logic combination of thedigital signals DCP<0> through DCP<2>, and is (111) at point-in-time T0,and is counted down toward (000) from point-in-time T1.

The binary numeral DCM is made up of the logic combination of thedigital signals DCM<0> through DCM<2>, and is (000) at point-in-time T0,and is counted up toward (111) when the binary numeral DCM reaches(000).

The signal LP is a signal synchronized with the signal L, wherein theleading-edge point-in-time from logic “L” to logic “H” is delayedaccording to the binary numeral DCP as compared to the signal L. Thedegree of delay thereof is the maximum when the binary numeral DCP is(111), and is the minimum when the binary numeral DCP is (000).

The signal LM is a signal synchronized with the signal L, wherein theleading-edge point-in-time from logic “L” to logic “H” is delayedaccording to the binary numeral DCM as compared to the signal L. Thedegree of delay thereof is the maximum when the binary numeral DCM is(000), and is the minimum when the binary numeral DCM is (111).

The signal OP is a signal indicating the comparison result of thepotential of the signal VIP and the potential of the signal VIM whenboth of the signals LP and LM are logic “H”. At point-in-time Tm+2, thelogic of the signal OP is logic “L”. As a result thereof, the count-upof the binary numeral DCM ends, and the values thereof are maintained.

FIGS. 5A and 5B are diagrams representing change in the signalpotentials of the signals OP and OM when the leading-edge point-in-timeof the signal LP or signal LM is delayed.

FIG. 5A is a diagram representing change in the signal potentials of thesignals OP and OM when the leading-edge period of the signal LM isfixed, and the leading-edge delay amount of the signal LP decreases.(VIP-VIM) indicates the voltage difference between the signals VIP andVIM, and (VIP-VIM) is 0. The signal L is synchronized with the reversedphase of the clock signal CK, and also the period of logic “H”, and theperiod of logic “L” are generally the same as with the clock signal CK.

The logic leading-edge delay amount of the signal LP varies according tothe magnitude of the binary numeral DCP made up of the digital signalsDCP<0> through DCP<2> as to the logic leading edge of the signal L. Thesignal OP is the output signal to be output from the latch unit 20 ofthe comparison circuit 10.

Now, let us say that the logic leading edge of the signal LP is delayedsuch as the waveform of the signal LP shown by a dotted line, and thesignals L and LM simultaneously logically rise.

In this case, the potential of the node A deteriorates toward apotential lower than the potential of the high-potential VDD powersource 60 by an amount corresponding to the multiplying of the potentialof the high-potential VDD power source 60 by the result obtained bydividing the on resistance of the P-type MOS transistor 21 by theresistance of the whole first current route, but the deterioration ofthe potential of the node B to be connected to the signal OP is littlesuch as the signal OP shown by a dotted line. This is becausehigh-potential VDD is applied to the node B by the P-type MOS transistor22 until the logic of the signal LP rises.

Accordingly, with the comparison circuit 10, the potential of the node Ais lower than the potential of the node B. As a result thereof, upon thelogic of the signal LP rising, the potential difference between thenodes A and B is expanded by the operation of the latch unit 20, and thepotential of the signal OM becomes equal to or smaller than apredetermined threshold from the potential VDD of the high-potential VDDpower source 60. As a result thereof, the logic of the signal OM isdetermined to be “L”, and the logic of the signal OP is determined to be“H”.

Next, upon the binary numeral DCP represented by the digital signalsDCP<0> through DCP<2> being counted down such as the waveform of thesignal LP shown by a solid line, the delay amount of the logic leadingedge of the signal LP decreases.

Thus, the logic of the signal LP rises at a stage wherein thedeterioration of the potential of the node A is small.

Now, VIP is equal to VIM, and accordingly, if we say that the propertiesof the on resistances of the N-type MOS transistors 31 and 32 of theinput unit 30 are equal as to gate voltage, the similar properties ofthe P-type MOS transistors 21 and 22 of the latch unit 20 are equal, andfurther the similar properties of the N-type MOS transistors 23 and 24of the latch unit 20 are equal, the potential difference between thenodes A and B is amplified as is due to the deterioration of thepotential of the node A.

However, in any of a case where the properties as to the gate voltageare arranged so that the on resistance of the N-type MOS transistor 31of the input unit 30 is higher than the on resistance of the N-type MOStransistor 32, a case where the properties are arranged so that the onresistance of the P-type MOS transistor 21 of the latch unit 20 is lowerthan the on resistance of the P-type MOS transistor 22, further a casewhere the properties are arranged so that the on resistance of theN-type MOS transistor 23 of the latch unit 20 is higher than the onresistance of the N-type MOS transistor 24, upon the logic of the signalLP rising, even if the potential of the node A deteriorates, thepotentials of the nodes A and B may be inverted. In this case, upon thepotential of the node B to be connected to the signal OP firstdeteriorating less than a threshold such as the signal OP shown by asolid line, the signal OP is determined to be logic “L”, and the logicof the signal OM is determined to be “H”.

However, the delay amount of the logic leading edge of the signal LP iskept great, whereby the potential of the node A, and the potential ofthe node B can be prevented from being inverted. In this case, the logicof the signal OM is determined to be “L”, and the logic of the signal OPis determined to be “H”.

FIG. 5B is a diagram representing change in the signal potentials of thesignals OP and OM when the leading-edge period of the signal LP isfixed, and the leading-edge delay amount of the signal LM increases.(VIP-VIM) indicates the voltage difference between the signals VIP andVIM, and (VIP-VIM) is 0. The signal L is synchronized with the reversedphase of the clock signal CK, and also the period of logic “H”, and theperiod of logic “L” are generally the same as with the clock signal CK.

The logic leading-edge delay amount of the signal LM varies according tothe magnitude of the binary numeral DCM made up of the digital signalsDCM<0> through DCM<2> as to the logic leading edge of the signal L. Thesignal OM is the output signal to be output from the latch unit 20 ofthe comparison circuit 10.

First, let us say that that the logic leading edge of the signal LM isnot delayed so much such as the waveform of the signal LM shown by adotted line, and the signals L and LP simultaneously logically rise.

In this case, the potential of the node B deteriorates toward apotential lower than the potential of the high-potential VDD powersource 60 by an amount corresponding to multiplying of the potential ofthe high-potential VDD power source 60 by the result obtained bydividing the on resistance of the P-type MOS transistor 22 by theresistance of the whole second current route, and the deterioration ofthe potential of the node A to be connected to the signal OM occurs suchas the signal OM shown by a dotted line.

Now, VIP is equal to VIM, and accordingly, if we say that the propertiesof the on resistances of the N-type MOS transistors 31 and 32 of theinput unit 30 are equal as to gate voltage, the similar properties ofthe P-type MOS transistors 21 and 22 of the latch unit 20 are equal, andfurther the similar properties of the N-type MOS transistors 23 and 24of the latch unit 20 are equal, the potential difference between thenodes A and B is amplified as is due to the deterioration of thepotential of the node B.

However, in any case of a case where the properties as to the gatevoltage are arranged so that the on resistance of the N-type MOStransistor 31 of the input unit 30 is lower than the on resistance ofthe N-type MOS transistor 32, a case where the properties are arrangedso that the on resistance of the P-type MOS transistor 21 of the latchunit 20 is higher than the on resistance of the P-type MOS transistor22, and further a case where the properties are arranged so that the onresistance of the N-type MOS transistor 23 of the latch unit 20 is lowerthan the on resistance of the N-type MOS transistor 24, upon the logicof the signal LM rising, even if the potential of the node Bdeteriorates, the potentials of the nodes A and B may be inverted suchas the signal OM indicated with a dotted line. In this case, the signalOP is determined to be logic “L” such as the signal OM shown by a dottedline, and the logic of the signal OM is determined to be “H” such as thesignal OP shown by a dotted line.

However, upon the logic leading-edge delay amount of the signal LMincreasing such as the signal LM shown by a solid line, thedeterioration of the potential of the node B to be connected to thesignal OP shown by a solid line becomes great, whereby the potential ofthe node A, and the potential of the node B is prevented from beinginverted. As a result thereof, according to the latch unit 30, the logicof the signal OM becomes logic “H”, and the logic of the signal OPbecomes logic “L”.

Thus, in any case of a case where the properties as to the gate voltagebetween the on resistance of the N-type MOS transistor 31 of the inputunit 30 and the on resistance of the N-type MOS transistor 32 differ, acase where the similar properties of the on resistance of the P-type MOStransistor 21 of the latch unit 20, and the on resistance of the P-typeMOS transistor 22 differ, and further a case where the similarproperties of the on resistance of the N-type MOS transistor 23 of thelatch unit 20, and the on resistance of the N-type MOS transistor 24differ, the logic leading-edge delay amount of the signal LM or signalLP is adjusted, whereby the comparison circuit 10 can be caused toexecute the same operation as in the case that the on resistanceproperties of each MOS resistor, and the transistor correspondingthereto are matched. As a result thereof, with the comparison circuit10, in the event of executing comparison between the voltage of thesignal VIP and the voltage of the signal VIM, error can be preventedfrom occurring due to the difference of the properties of the MOStransistors making up the comparison circuit 10, or the amplificationand held properties of the latch unit 20.

Thus, the comparison circuit 10 according to the first embodiment is acomparison circuit including an input unit 30 made up of a first MOStransistor (N-type MOS transistor 31) configured to receive a firstsignal at the gate electrode, and a second MOS transistor configured toreceive a second signal at the gate; a latch circuit 20 configured toamplify potential difference between a first current route where thecurrent is controlled by the first MOS transistor according to thevoltage of the first signal, and a second current route where thecurrent is controlled by the second MOS transistor according to thevoltage of the second signal; a comparative operation control unitincluding a first switch configured to execute supply or blocking ofsupply of high-potential VDD to the drain of the first MOS transistor bya third current route different from the first current route, and asecond switch configured to execute supply or blocking of supply of thehigh-potential VDD to the drain of the second MOS transistor by a fourthcurrent route different from the second current route, and a thirdswitch configured to execute supply or blocking of supply of a lowpotential to the first current route and second current route; and acomparative operation setting unit configured to set the period ofsupply or blocking of supply of the first switch, second switch, andthird switch.

The comparative operation setting unit (comparative operation settingunit 50) includes a delay circuit configured to determine a periodbetween a blocking period of supply of high-potential VDD by the firstswitch, and a blocking period of supply of high-potential VDD by thesecond switch, and a setting circuit configured to execute setting of aperiod.

Thus, with the comparison circuit 10 according to the first embodiment,timing of supply or blocking of supply of high-potential VDD to thethird current route or the fourth current route by the first switch andthe second switch, whereby the potentials of the first current route andthe second current route before comparison can be controlled.

As a result thereof, the potentials of the first current route and thesecond current route before comparison are controlled, wherebycomparison error along with difference of the properties relating to thegate voltage of the on resistances of the first MOS transistor and thesecond MOS transistor which receive the input signals VIP and VIM can becontrolled.

Second Embodiment

FIG. 6 is a circuit diagram illustrating a comparison circuit 100according to the second embodiment. The comparison circuit 100 includesa P-type MOS transistor latch unit 120, an input 130, an N-type MOStransistor latch unit 125, a comparative operation control circuit 140,and a comparative operation setting circuit 150.

The comparative operation control unit 140 includes P-type MOStransistors 143, 144, 145, and 146, and N-type MOS transistors 141, 142,147, and 148.

The N-type MOS transistor 141 has a drain to be connected to the sourceof an N-type MOS transistor 131 of the input 130, a source to beconnected to the ground VSS 70, and a gate for receiving a signal LM2.

The N-type MOS transistor 142 has a drain to be connected to the sourceof an N-type MOS transistor 132 of the input 130, a source to beconnected to the ground VSS 70, and a gate for receiving a signal LP2.

The N-type MOS transistor 141 supplies the ground potential from theground VSS 70 to the input unit 130 when the logic of the signal LM2 is“H”, and blocks supply of the ground potential from the ground VSS 70 tothe input unit 130 when the logic is “H”. The N-type MOS transistor 141serves as a switch for connecting or blocking between the input unit 130and the ground VSS 70.

The N-type MOS transistor 142 supplies the ground potential from theground VSS 70 to the input unit 130 when the logic of the signal LP2 is“H”, and blocks supply of the ground potential from the ground VSS 70 tothe input unit 130 when the logic is “L”. The N-type MOS transistor 142serves as a switch for connecting or blocking between the input unit 130and the ground VSS 70.

The P-type MOS transistor 143 has a source to be connected to thehigh-potential VDD power source 60, a drain to be connected to the drainof the P-type MOS transistor 121 of the P-type MOS transistor latch unit120, and a gate for receiving a signal LM1.

The P-type MOS transistor 143 blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the P-type MOS transistorlatch unit 120 when the logic of the signal LM1 is “H”, and supplies thehigh-potential VDD from the high-potential VDD power source 60 to theP-type MOS transistor latch unit 120 when the logic is “L”. The P-typeMOS transistor 143 serves as a switch for connecting or blocking betweenthe P-type MOS transistor latch unit 120 and the high-potential VDDpower source 60.

The P-type MOS transistor 144 has a source to be connected to thehigh-potential VDD power source 60, a drain to be connected to the drainof the N-type MOS transistor 126 of the N-type MOS transistor latch unit125, and a gate for receiving a signal LM0.

The P-type MOS transistor 144 blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the N-type MOS transistorlatch unit 125 when the logic of the signal LM0 is “H”, and supplies thehigh-potential VDD from the high-potential VDD power source 60 to theN-type MOS transistor latch unit 125 when the logic is “L”. The P-typeMOS transistor 144 serves as a switch for connecting or blocking betweenthe N-type MOS transistor latch unit 125 and the high-potential VDDpower source 60.

The P-type MOS transistor 145 has a source to be connected to thehigh-potential VDD power source 60, a drain to be connected to the drainof the N-type MOS transistor 127 of the N-type MOS transistor latch unit125, and a gate for receiving a signal LP0.

The P-type MOS transistor 145 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the N-type MOS transistor latchunit 125 when the logic of the signal LM0 is “H”, and blocks supply ofthe high-potential VDD from the high-potential VDD power source 60 tothe N-type MOS transistor latch unit 125 when the logic is “L”. TheP-type MOS transistor 145 serves as a switch for connecting or blockingbetween the N-type MOS transistor latch unit 125 and the high-potentialVDD power source 60.

The P-type MOS transistor 146 has a source to be connected to thehigh-potential VDD power source 60, a drain to be connected to the drainof the P-type MOS transistor 122 of the P-type MOS transistor latch unit120, and a gate for receiving a signal LP1.

The P-type MOS transistor 146 blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the P-type MOS transistorlatch unit 120 when the logic of the signal LP1 is “H”, and supplies thehigh-potential VDD from the high-potential VDD power source 60 to theP-type MOS transistor latch unit 120 when the logic is “L”. The P-typeMOS transistor 146 serves as a switch for connecting or blocking betweenthe P-type MOS transistor latch unit 120 and the high-potential VDDpower source 60.

The N-type MOS transistor 147 has a drain to be connected to the drainof the P-type MOS transistor 121 of the P-type MOS transistor latch unit120, a source to be connected to the drain of the N-type MOS transistor126 of the N-type MOS transistor latch unit 125, and a gate forreceiving a signal LM2.

The N-type MOS transistor 148 has a drain to be connected to the drainof the P-type MOS transistor 122 of the P-type MOS transistor latch unit120, a source to be connected to the drain of the N-type MOS transistor127 of the N-type MOS transistor latch unit 125, and a gate forreceiving a signal LP2.

The input unit 130 includes the N-type MOS transistors 131 and 132. TheN-type MOS transistor 131 has a drain to be connected to the source ofthe N-type MOS transistor 126 of the N-type MOS transistor latch unit125, a source to be connected to the drain of the N-type MOS transistor141 of the comparative operation control circuit 140, and a gate forreceiving the input signal VIP. The on resistance value of the N-typeMOS transistor 131 varies depending on the potential of the input signalVIP.

The N-type MOS transistor 132 has a drain to be connected to the sourceof the N-type MOS transistor 127 of the N-type MOS transistor latch unit125, a source to be connected to the drain of the N-type MOS transistor142 of the comparative operation control circuit 140, and a gate forreceiving the signal VIM. The on resistance value of the N-type MOStransistor 132 varies depending on the potential of the signal VIM.

The P-type MOS transistor latch unit 120 includes the P-type MOStransistors 121 and 122. The N-type MOS transistor latch unit 125includes the N-type MOS transistors 126 and 127.

The P-type MOS transistor 121 has a drain to be connected to the drainof the N-type MOS transistor 147, a gate to be connected to the drain ofthe P-type MOS transistor 122, and a source to be connected to thehigh-potential VDD power source 60.

The P-type MOS transistor 122 has a drain to be connected to the drainof the N-type MOS transistor 148, a gate to be connected to the drain ofthe P-type MOS transistor 121, and a source to be connected to thehigh-potential VDD power source 60.

The N-type MOS transistor 126 has a drain to be connected to the sourceof the N-type MOS transistor 147, a gate to be connected to the drain ofthe N-type MOS transistor 127, and a source to be connected to the drainof the N-type MOS transistor 131 of the input unit 130.

The N-type MOS transistor 127 has a drain to be connected to the sourceof the N-type MOS transistor 148, a gate to be connected to the drain ofthe N-type MOS transistor 126, and a source to be connected to the drainof the N-type MOS transistor 132 of the input unit 130.

The output signals OM and OP are output from the P-type MOS transistorlatch unit 120. The output signal OM is connected to the node A betweenthe drain of the P-type MOS transistor 121 and the drain of the N-typeMOS transistor 147. The output signal OP is connected to the node Bbetween the drain of the P-type MOS transistor 122 and the drain of theN-type MOS transistor 148.

The gate of the P-type MOS transistor 121 of the P-type MOS transistorlatch unit 120 and the drain of the P-type MOS transistor 122 areconnected to the node B, and the gate of the P-type MOS transistor 122and the drain of the P-type MOS transistor 121 are connected to the nodeA. That is to say, the P-type MOS transistor 121 and the P-type MOStransistor 122 are connected crosswise to the nodes A and B, andaccordingly, the P-type MOS transistor 121 and the P-type MOS transistor122 amplify potential difference between the nodes A and B.

The gate of the N-type MOS transistor 126 and the drain of the N-typeMOS transistor 127 of the N-type MOS transistor latch unit 125 areconnected to the drain (node D) of the N-type MOS Transistor 148, andthe gate of the N-type MOS transistor 127 and the drain of the N-typeMOS transistor 126 are connected to the drain (node C) of the N-type MOStransistor 147. That is to say, the N-type MOS transistor 126 and theN-type MOS transistor 127 are connected crosswise to the nodes C and D,and accordingly, the N-type MOS transistor 126 and the N-type MOStransistor 127 amplify potential difference between the nodes C and D.

Thus, the P-type MOS transistor 121 of the P-type MOS transistor latchunit 120, the N-type MOS transistor 147, the N-type MOS transistor 126,the N-type MOS transistor 131, and the N-type MOS transistor 141 areserially connected between the high-potential VDD power source 60 andthe ground VSS 70, and make up a first current route including the nodesA and C. The P-type MOS transistor 122 of the P-type MOS transistorlatch unit 120, the N-type MOS transistor 148, the N-type MOS transistor127, the N-type MOS transistor 132, and the N-type MOS transistor 142are serially connected between the high-potential VDD power source 60and the ground VSS 70, and make up a second current route including thenodes B and D.

Therefore, when the signals LM0, LM1, LM2, LP0, LP1, and LP2 are “H”,supply of the high-potential VDD to the P-type MOS transistor latch unit120 and the N-type MOS transistor latch unit 125 is blocked by theP-type MOS transistors 143, 144, 145, and 146, and the ground potentialVSS is supplied to the input unit 130, the P-type MOS transistor latchunit 120, and the N-type MOS transistor latch unit 125 by the N-type MOStransistors 141 and 142.

In the above case, the on resistance of the N-type MOS transistor 131varies according to the potential of the input signal VIP, andaccordingly, upon the potential of the input signal VIP decreasing, thepotentials of the nodes A and C increase, and the on resistance of theN-type MOS transistor 132 varies according to the potential of the inputsignal VIM, and accordingly, upon the potential of the input signal VIMincreasing contrary to the input signal VIP, the potentials of the nodesB and D decrease. On the other hand, conversely, upon the potential ofthe input signal VIP increasing, the potentials of the nodes A and Cdecrease, and upon the potential of the input signal VIM decreasing, thepotentials of the nodes B and D increase.

Note that, when the signals LM0, LM1, LM2, LP0, LP1, and LP2 are “L”,the high-potential VDD is supplied to the P-type MOS transistor latchunit 120, and the N-type MOS transistor latch unit 125, and the inputunit 130 by the P-type MOS transistors 143, 144, 145, and 146, and thesupply of the ground potential to the input unit 130 is blocked by theN-type MOS transistors 141 and 142.

As a result thereof, potential difference between the nodes A and B is0, or almost eliminated.

The comparative operation setting circuit 150 is made up of a circuitsimilar to the comparative operation setting circuit 50 according to thefirst embodiment for driving the signals LM0 and LP0, a circuit similarto the comparative operation setting circuit 50 for driving the signalsLM1 and LP1, and a circuit similar to the comparative operation settingcircuit 50 for driving the signals LM2 and LP2. Also, the circuitssimilar to the comparative operation setting circuit 50 include a delaycircuit 151, and a logic circuit 156 for controlling the delay circuit151. The delay circuit 151 and the logic circuit 156 are the samecircuits as the delay circuit 51 and the logic circuit 56 according tothe first embodiment. Now, it goes without saying that delay between thesignals LM0 and LP0, delay between the signals LM1 and LP1, and delaybetween the signals LM2 and LP2 may be set to the same delay amount, ormay be set separately. It is further needless to say that an arrangementmay be made where just one of the delays is set.

Thus, the comparison circuit 100 according to the second embodimentincludes an input unit (130) made up of a first MOS transistor (N-typeMOS transistor 131) for receiving a first signal (signal VIP) at thegate electrode, and a second MOS transistor (N-type MOS transistor 132)for receiving a second signal (signal VIM) at the gate electrode; alatch circuit (P-type MOS transistor latch circuit 120, N-type MOStransistor latch circuit 125) for amplifying potential differencebetween a node within a first current route where an electric current iscontrolled by the first MOS transistor according to the voltage of thefirst signal, and a node within a second current route where an electriccurrent is controlled by the second MOS transistor according to thevoltage of the second signal; a comparative operation setting unit(comparative operation setting circuit 150) including a first switch(P-type MOS transistor 144) for executing supply or blocking of supplyof the high-potential VDD to the drain of the first MOS transistor by athird current route different from the first current route, a secondswitch (P-type MOS transistor 145) for executing supply or blocking ofsupply of the high-potential VDD to the drain of the second MOStransistor by a fourth current route different from the second currentroute, and a third switch (N-type MOS transistors 141 and 147) and afourth switch (N-type MOS transistors 142 and 148) which execute supplyor blocking of supply of a low potential to the first current route andthe second current route; a fifth switch (P-type MOS transistor 143) tobe connected to the latch circuit; a sixth switch (P-type MOS transistor146) to be connected to the latch circuit; and a comparative operationcontrol unit (comparative operation control circuit 140) for controllingsupply or blocking of the first through sixth switches.

Thus, supply or blocking of any of the first through sixth switches iscontrolled, whereby the potentials of the first current route and thesecond current route before start of comparison can also be controlledat the comparison circuit 100 according to the second embodiment.

As a result thereof, the potentials of the first current route and thesecond current route before start of comparison are controlled, wherebycomparison error along with different properties relating to the gatevoltages of on resistances of the first and second MOS transistors whichreceive input signals VIP and VIM can be controlled. Note that it goeswithout saying that comparison error can be controlled by controllingany of the first through sixth switches.

Further, with the comparison circuit 100 according to the secondembodiment, the latch circuit unit is made up of the N-type MOStransistor latch circuit 120 and the P-type MOS transistor latch circuit125, and accordingly, control of supply or blocking of supply of thehigh-potential VDD to the latch circuits is ensured by providing thefifth switch (P-type MOS transistor 143) and the sixth switch (P-typeMOS transistor 146) to be connected to the latch circuits.

Third Embodiment

FIG. 7 is a circuit diagram illustrating a comparison circuit 200according to the third embodiment. The comparison circuit 200 includes aP-type MOS transistor latch unit 220, an input unit 230, an N-type MOStransistor latch unit 225, a comparative operation control circuit 240,and a comparative operation setting circuit 250.

The comparative operation control circuit 240 includes N-type MOStransistors 243, 244, 245, and 246, and P-type MOS transistors 241, 242,247, and 248.

The P-type MOS transistor 241 has a drain to be connected to the sourceof a P-type MOS transistor 231 of the input unit 230, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LM2.

The P-type MOS transistor 242 has a drain to be connected to the sourceof a P-type MOS transistor 232 of the input unit 230, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LP2.

The P-type MOS transistor 241 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 230 when the logicof the signal LM2 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 230 whenthe logic is “H”. The P-type MOS transistor 241 serves as a switch forconnecting or blocking between the input unit 230 and the high-potentialVDD power source 60.

The P-type MOS transistor 242 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 230 when the logicof the signal LP2 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 230 whenthe logic is “H”. The P-type MOS transistor 242 serves as a switch forconnecting or blocking between the input unit 230 and the high-potentialVDD power source 60.

The N-type MOS transistor 243 has a source to be connected to a groundVSS power source 70, a drain to be connected to the drain of an N-typeMOS transistor 226 of the N-type MOS transistor latch unit 225, and agate for receiving the signal LM1.

The N-type MOS transistor 243 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the N-type MOS transistor latchunit 225 when the logic of the signal LM1 is “H”, and blocks supply ofthe high-potential VDD from the high-potential VDD power source 60 tothe N-type MOS transistor latch unit 225 when the logic is “L”. TheN-type MOS transistor 243 serves as a switch for connecting or blockingbetween the N-type MOS transistor latch unit 225 and the high-potentialVDD power source 60.

The N-type MOS transistor 244 has a source to be connected to the groundVSS power source 70, a drain to be connected to the source of a P-typeMOS transistor 221 of the P-type MOS transistor latch unit 220, and agate for receiving the signal LM0.

The N-type MOS transistor 244 supplies the ground VSS potential from theground VSS power source 70 to the P-type MOS transistor latch unit 220when the logic of the signal LM0 is “H”, and blocks supply of the groundVSS potential from the ground VSS power source 70 to the P-type MOStransistor latch unit 220 when the logic is “L”. The N-type MOStransistor 244 serves as a switch for connecting or blocking between theP-type MOS transistor latch unit 220 and the ground VSS power source 70.

The N-type MOS transistor 245 has a source to be connected to the groundVSS power source 70, a drain to be connected to the source of a P-typeMOS transistor 222 of the P-type MOS transistor latch unit 220, and agate for receiving the signal LP0.

The N-type MOS transistor 245 supplies the ground VSS potential from theground VSS power source 70 to the P-type MOS transistor latch unit 220when the logic of the signal LP0 is “H”, and blocks supply of the groundVSS potential from the ground VSS power source 70 to the P-type MOStransistor latch unit 220 when the logic is “L”. The N-type MOStransistor 245 serves as a switch for connecting or blocking between theP-type MOS transistor latch unit 220 and the ground VSS power source 70.

The N-type MOS transistor 246 has a source to be connected to the groundVSS power source 70, a drain to be connected to the drain of an N-typeMOS transistor 227 of the N-type MOS transistor latch unit 225, and agate for receiving the signal LP1.

The N-type MOS transistor 246 supplies the ground VSS from the groundVSS power source 70 to the N-type MOS transistor latch unit 225 when thelogic of the signal LP1 is “H”, and blocks supply of the ground VSS fromthe ground VSS power source 70 to the N-type MOS transistor latch unit225 when the logic is “L”. The N-type MOS transistor 246 serves as aswitch for connecting or blocking between the N-type MOS transistorlatch unit 225 and the ground VSS power source 70.

The P-type MOS transistor 247 has a source to be connected to the drainof the P-type MOS transistor 221 of the P-type MOS transistor latch unit220, a drain to be connected to the drain of the N-type MOS transistor226 of the N-type MOS transistor latch unit 225, and a gate forreceiving the signal LM2.

The P-type MOS transistor 248 has a source to be connected to the drainof the P-type MOS transistor 222 of the P-type MOS transistor latch unit220, a drain to be connected to the drain of the N-type MOS transistor227 of the N-type MOS transistor latch unit 225, and a gate forreceiving the signal LP2.

The input unit 230 includes the P-type MOS transistors 231 and 232. TheP-type MOS transistor 231 has a drain to be connected to the source ofthe P-type MOS transistor 221 of the P-type MOS transistor latch unit220, a source to be connected to the drain of the P-type MOS transistor241 of the comparative operation control circuit 240, and a gate forreceiving an input signal VIP. The on resistance value of the P-type MOStransistor 231 varies according to the potential of the input signalVIP.

The P-type MOS transistor 232 has a drain to be connected to the sourceof the P-type MOS transistor 222 of the P-type MOS transistor latch unit220, a source to be connected to the drain of the P-type MOS transistor242 of the comparative operation control circuit 240, and a gate forreceiving a signal VIM. The on resistance value of the P-type MOStransistor 232 varies according to the potential of the signal VIM.

The P-type MOS transistor latch unit 220 includes the P-type MOStransistors 221 and 222. The N-type MOS transistor latch unit 225includes the N-type MOS transistors 226 and 227.

The P-type MOS transistor 221 has a drain to be connected to the sourceof the P-type MOS transistor 247, a source to be connected to the drainof the P-type MOS transistor 231, and a gate to be connected to thedrain of the P-type MOS transistor 222.

The P-type MOS transistor 222 has a drain to be connected to the sourceof the P-type MOS transistor 248, a source to be connected to the drainof the P-type MOS transistor 232, and a gate to be connected to thedrain of the P-type MOS transistor 221.

The N-type MOS transistor 226 has a drain to be connected to the drainof the P-type MOS transistor 247, a gate to be connected to the drain ofthe N-type MOS transistor 227, and a source to be connected to theground VSS 70.

The N-type MOS transistor 227 has a drain to be connected to the drainof the P-type MOS transistor 248, a gate to be connected to the drain ofthe N-type MOS transistor 226, and a source to be connected to theground VSS 70.

The output signals OM and OP are output from the N-type MOS transistorlatch unit 225. The output signal OM is connected to the node A betweenthe drain of the N-type MOS transistor 226 and the drain of the P-typeMOS transistor 247. The output signal OP is connected to the node Bbetween the drain of the N-type MOS transistor 227 and the drain of theP-type MOS transistor 248.

The gate of the N-type MOS transistor 226 of the N-type MOS transistorlatch unit 225 and the drain of the N-type MOS transistor 227 areconnected to the node B, and the gate of the N-type MOS transistor 227and the drain of the N-type MOS transistor 226 are connected to the nodeA. That is to say, the N-type MOS transistor 226 and the N-type MOStransistor 227 are connected crosswise to the nodes A and B, andaccordingly, the N-type MOS transistor 226 and the N-type MOS transistor227 amplify potential difference between the nodes A and B.

The gate of the P-type MOS transistor 221 and the drain of the P-typeMOS transistor 222 of the P-type MOS transistor latch unit 220 areconnected to the source (node D) of the P-type MOS Transistor 248, andthe gate of the P-type MOS transistor 222 and the drain of the P-typeMOS transistor 221 are connected to the source (node C) of the P-typeMOS transistor 247. That is to say, the P-type MOS transistor 226 andthe P-type MOS transistor 227 are connected crosswise to the nodes C andD, and accordingly, the P-type MOS transistor 221 and the P-type MOStransistor 222 amplify potential difference between the nodes C and D.

Thus, the P-type MOS transistor 221 of the P-type MOS transistor latchunit 220, the P-type MOS transistor 247, the N-type MOS transistor 226,the P-type MOS transistor 231, and the P-type MOS transistor 241 areserially connected between the high-potential VDD power source 60 andthe ground VSS 70, and make up a first current route including the nodesA and C. The P-type MOS transistor 222 of the P-type MOS transistorlatch unit 220, the P-type MOS transistor 248, the N-type MOS transistor227, the P-type MOS transistor 232, and the P-type MOS transistor 242are serially connected between the high-potential VDD power source 60and the ground VSS 70, and make up a second current route including thenodes B and D.

Therefore, when the signals LM0, LM1, LM2, LP0, LP1, and LP2 are “L”,supply of the ground VSS to the P-type MOS transistor latch unit 220 andthe N-type MOS transistor latch unit 225 is blocked by the N-type MOStransistors 243, 244, 245, and 246, and the high-potential VDD issupplied to the input unit 230, the P-type MOS transistor latch unit220, and the N-type MOS transistor latch unit 225 by the P-type MOStransistors 241 and 242.

In the above case, the on resistance of the P-type MOS transistor 231varies according to the potential of the input signal VIP, andaccordingly, upon the potential of the input signal VIP decreasing, thepotentials of the nodes A and C decrease, and the on resistance of theN-type MOS transistor 232 varies according to the potential of the inputsignal VIM, and accordingly, upon the potential of the input signal VIMincreasing contrary to the input signal VIP, the potentials of the nodesB and D increase. On the other hand, conversely, upon the potential ofthe input signal VIP increasing, the potentials of the nodes A and Cincrease, and upon the potential of the input signal VIM decreasing, thepotentials of the nodes B and D decrease.

Note that, when the signals LM0, LM1, LM2, LP0, LP1, and LP2 are “H”,the ground VSS is supplied to the P-type MOS transistor latch unit 220,the N-type MOS transistor latch unit 225, and the input unit 230 by theN-type MOS transistors 243, 244, 245, and 246, and the supply of thehigh-potential VDD to the input unit 230 is blocked by the P-type MOStransistors 241 and 242.

As a result thereof, potential difference between the nodes A and B is0, or almost eliminated.

The comparative operation setting circuit 250 is made up of a circuitsimilar to the comparative operation setting circuit 50 according to thefirst embodiment for driving the signals LM0 and LP0, a circuit similarto the comparative operation setting circuit 50 for driving the signalsLM1 and LP1, and a circuit similar to the comparative operation settingcircuit 50 for driving the signals LM2 and LP2. Also, the circuitssimilar to the comparative operation setting circuit 50 include a delaycircuit 251, and a logic circuit 256 for controlling the delay circuit251. Also, the delay circuit 251 and the logic circuit 256 are the samecircuits as the delay circuit 51 and the logic circuit 56. Now, it goeswithout saying that delay between the signals LM0 and LP0, delay betweenthe signals LM1 and LP1, and delay between the signals LM2 and LP2 maybe set to the same delay amount, or may be set separately. It is furtherneedless to say that an arrangement may be made where just one of thedelays is set.

Thus, the comparison circuit 200 according to the third embodimentincludes an input unit (230) made up of a first MOS transistor (P-typeMOS transistor 231) for receiving a first signal (signal VIP) at thegate electrode, and a second MOS transistor (P-type MOS transistor 232)for receiving a second signal (signal VIM) at the gate electrode; alatch circuit (P-type MOS transistor latch circuit 220, N-type MOStransistor latch circuit 225) for amplifying potential differencebetween a first current route where an electric current is controlled bythe first MOS transistor according to the voltage of the first signal,and a second current route where an electric current is controlled bythe second MOS transistor according to the voltage of the second signal;a comparative operation setting unit (comparative operation settingcircuit 250) including a first switch (N-type MOS transistor 244) forexecuting supply or blocking of supply of the ground VSS to the drain ofthe first MOS transistor by a third current route different from thefirst current route, a second switch (N-type MOS transistor 245) forexecuting supply or blocking of supply of the ground VSS to the drain ofthe second MOS transistor by a fourth current route different from thesecond current route, and a third switch (P-type MOS transistors 241 and247) and a fourth switch (P-type MOS transistors 242 and 248) whichexecute supply or blocking of supply of the high-potential VDD to thefirst current route and the second current route; a fifth switch (N-typeMOS transistor 243) to be connected to the latch circuit; a sixth switch(N-type MOS transistor 246) to be connected to the latch circuit; and acomparative operation control unit (comparative operation controlcircuit 240) for controlling supply or blocking of the first throughsixth switches.

Thus, supply or blocking timing of any of the first through sixthswitches is controlled, whereby the potentials of the first currentroute and the second current route before start of comparison can alsobe controlled.

As a result thereof, the potentials of the first current route and thesecond current route before start of comparison are controlled, wherebycomparison error along with different properties relating to the gatevoltages of on resistances of the first and second MOS transistors whichreceive input signals VIP and VIM can be controlled. Note that it goeswithout saying that comparison error can be controlled by controllingany of the first through sixth switches.

Further, with the comparison circuit 200 according to the thirdembodiment, the latch circuit portion is made up of the N-type MOStransistor latch circuit 220 and the P-type MOS transistor latch circuit225, and accordingly, control of supply or blocking of supply of theground VSS to the latch circuits is ensured by providing the fifthswitch (N-type MOS transistor 243) and the sixth switch (N-type MOStransistor 246) to be connected to the latch circuits.

Fourth Embodiment

FIG. 8 is a circuit diagram illustrating a comparison circuit 300according to the fourth embodiment. The comparison circuit 300 includesa P-type MOS transistor latch unit 320, an input unit 330, an N-type MOStransistor latch unit 325, a comparative operation control circuit 340,and a comparative operation setting circuit 350.

The comparative operation control circuit 340 includes N-type MOStransistors 343 and 344, and P-type MOS transistors 341 and 342.

The P-type MOS transistor 341 has a drain to be connected to the drainof an N-type MOS transistor 331 of the input unit 330, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LM1.

Note that the drain of the P-type MOS transistor 341, the drain of theN-type MOS transistor 331 of the input unit 330, the drain of the P-typeMOS transistor 321 of the P-type MOS transistor latch unit 320, and thedrain of the N-type MOS transistor 326 of the N-type MOS transistorlatch unit 325 are all connected to the node A within the comparisoncircuit 300. The comparison circuit 300 outputs an output signal OM fromthe node A.

The P-type MOS transistor 342 has a drain to be connected to the drainof an N-type MOS transistor 332 of the input unit 330, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LP1.

Note that the drain of the P-type MOS transistor 342, the drain of theN-type MOS transistor 332 of the input unit 330, the drain of the P-typeMOS transistor 322 of the P-type MOS transistor latch unit 320, and thedrain of the N-type MOS transistor 327 of the N-type MOS transistorlatch unit 325 are all connected to the node B within the comparisoncircuit 300. The comparison circuit 300 outputs an output signal OP fromthe node B.

The P-type MOS transistor 341 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 330 when the logicof the signal LM1 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 330 whenthe logic is “H”. The P-type MOS transistor 341 serves as a switch forconnecting or blocking between the input unit 330 and the high-potentialVDD power source 60.

The P-type MOS transistor 342 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 330 when the logicof the signal LP1 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 330 whenthe logic is “H”. The P-type MOS transistor 342 serves as a switch forconnecting or blocking between the input unit 330 and the high-potentialVDD power source 60.

The N-type MOS transistor 343 has a source to be connected to the groundVSS power source 70, a drain to be connected to the source of the N-typeMOS transistor 331 of the input unit 330, and a gate for receiving asignal LM0.

The N-type MOS transistor 343 supplies the ground VSS from the groundVSS power source 70 to the input unit 330 when the logic of the signalLM0 is “H”, and blocks supply of the ground VSS from the ground VSSpower source 70 to the input unit 330 when the logic is “L”. The N-typeMOS transistor 343 serves as a switch for connecting or blocking betweenthe input unit 330 and the ground VSS power source 70.

The N-type MOS transistor 344 has a source to be connected to the groundVSS power source 70, a drain to be connected to the source of the N-typeMOS transistor 332 of the input unit 330, and a gate for receiving asignal LP0.

The N-type MOS transistor 344 supplies the ground VSS from the groundVSS power source 70 to the input unit 330 when the logic of the signalLP0 is “H”, and blocks supply of the ground VSS from the ground VSSpower source 70 to the input unit 330 when the logic is “L”. The N-typeMOS transistor 344 serves as a switch for connecting or blocking betweenthe input unit 330 and the ground VSS power source 70.

The input unit 330 includes the N-type MOS transistors 331 and 332. TheN-type MOS transistor 331 has a drain to be connected to the drain ofthe P-type MOS transistor 321 of the P-type MOS transistor latch unit320, a source to be connected to the drain of the N-type MOS transistor343 of the comparative operation control circuit 340, and a gate forreceiving an input signal VIP. The on resistance value of the N-type MOStransistor 331 varies according to the potential of the input signalVIP.

The N-type MOS transistor 332 has a drain to be connected to the drainof the P-type MOS transistor 322 of the P-type MOS transistor latch unit320, a source to be connected to the drain of the N-type MOS transistor344 of the comparative operation control circuit 340, and a gate forreceiving a signal VIM. The on resistance value of the N-type MOStransistor 332 varies according to the potential of the signal VIM.

The P-type MOS transistor latch unit 320 includes the P-type MOStransistors 321 and 322. The N-type MOS transistor latch unit 325includes the N-type MOS transistors 326 and 327.

The P-type MOS transistor 321 has a drain to be connected to the drainof the N-type MOS transistor 326, a source to be connected to thehigh-potential VDD power source 60, and a gate to be connected to thedrain of the P-type MOS transistor 322.

The P-type MOS transistor 322 has a drain to be connected to the drainof the N-type MOS transistor 327, a source to be connected to thehigh-potential VDD power source 60, and a gate to be connected to thedrain of the P-type MOS transistor 321.

The N-type MOS transistor 326 has a drain to be connected to the drainof the P-type MOS transistor 321, a gate to be connected to the drain ofthe N-type MOS transistor 327, and a source to be connected to theground VSS 70.

The N-type MOS transistor 327 has a drain to be connected to the drainof the P-type MOS transistor 322, a gate to be connected to the drain ofthe N-type MOS transistor 326, and a source to be connected to theground VSS 70.

The output signals OM and OP are output from the N-type MOS transistorlatch unit 325. The output signal OM is connected to the node A betweenthe drain of the N-type MOS transistor 326 and the drain of the P-typeMOS transistor 321. The output signal OP is connected to the node Bbetween the drain of the N-type MOS transistor 327 and the drain of theP-type MOS transistor 322.

The gate of the N-type MOS transistor 326 of the N-type MOS transistorlatch unit 325 and the drain of the N-type MOS transistor 327 areconnected to the node B, and the gate of the N-type MOS transistor 327and the drain of the N-type MOS transistor 326 are connected to the nodeA. That is to say, the N-type MOS transistor 326 and the N-type MOStransistor 327 are connected crosswise to the nodes A and B, andaccordingly, the N-type MOS transistor 326 and the N-type MOS transistor327 amplify potential difference between the nodes A and B.

Thus, the P-type MOS transistor 321 of the P-type MOS transistor latchunit 320, the N-type MOS transistor 331, and the N-type MOS transistor343 are serially connected between the high-potential VDD power source60 and the ground VSS 70, and make up a first current route includingthe node A. The P-type MOS transistor 322 of the P-type MOS transistorlatch unit 320, the P-type MOS transistor 332, and the N-type MOStransistor 344 are serially connected between the high-potential VDDpower source 60 and the ground VSS 70, and make up a second currentroute including the node B.

Therefore, when the signals LM0, LM1, LP0, and LP1 are “H”, supply ofthe high-potential VDD to the P-type MOS transistor latch unit 320 andthe N-type MOS transistor latch unit 325 is blocked by the P-type MOStransistors 341 and 342, and the ground potential VSS is supplied to theinput unit 330, and the P-type MOS transistor latch unit 320 by theN-type MOS transistors 343 and 344.

In the above case, the on resistance of the P-type MOS transistor 331varies according to the potential of the input signal VIP, andaccordingly, upon the potential of the input signal VIP decreasing, thepotential of the node A increases, and the on resistance of the N-typeMOS transistor 332 varies according to the potential of the input signalVIM, and accordingly, upon the potential of the input signal VIMincreasing contrary to the input signal VIP, the potential of the node Bdecreases. On the other hand, conversely, upon the potential of theinput signal VIP increasing, the potential of the node A decreases, andupon the potential of the input signal VIM decreasing, the potential ofthe node B increases.

Note that, when the signals LM0, LM1, LP0, and LP1 are “L”, thehigh-potential VDD is supplied to the P-type MOS transistor latch unit320, and the N-type MOS transistor latch unit 325, and the input unit330 by the P-type MOS transistors 341 and 342, and the supply of theground VSS to the input unit 330 is blocked by the N-type MOStransistors 343 and 344.

As a result thereof, potential difference between the nodes A and B is0, or almost eliminated.

The comparative operation setting circuit 350 is made up of a circuitsimilar to the comparative operation setting circuit 50 according to thefirst embodiment for driving the signals LM0 and LP0, and a circuitsimilar to the comparative operation setting circuit 50 for driving thesignals LM1 and LP1. The circuits similar to the comparative operationsetting circuit 50 include a delay circuit 351, and a logic circuit 356for controlling the delay circuit 351. The delay circuit 351 and thelogic circuit 356 are the same circuits as the delay circuit 51 and thelogic circuit 56. Now, it goes without saying that delay between thesignals LM0 and LP0, and delay between the signals LM1 and LP1 may beset to the same delay amount, or may be set separately. It is furtherneedless to say that an arrangement may be made where just one of thedelays is set.

Thus, the comparison circuit 300 according to the fourth embodimentincludes an input unit (330) made up of a first MOS transistor (N-typeMOS transistor 331) for receiving a first signal (signal VIP) at thegate electrode, and a second MOS transistor (N-type MOS transistor 332)for receiving a second signal (signal VIM) at the gate electrode; alatch circuit (P-type MOS transistor latch circuit 320, N-type MOStransistor latch circuit 325) for amplifying potential differencebetween a first current route where an electric current is controlled bythe first MOS transistor according to the voltage of the first signal,and a second current route where an electric current is controlled bythe second MOS transistor according to the voltage of the second signal;a comparative operation setting unit (comparative operation settingcircuit 350) including a first switch (P-type MOS transistor 341) forexecuting supply or blocking of supply of the high-potential VDD to thedrain of the first MOS transistor by a third current route differentfrom the first current route, a second switch (P-type MOS transistor342) for executing supply or blocking of supply of the high-potentialVDD to the drain of the second MOS transistor by a fourth current routedifferent from the second current route, and a third switch (N-type MOStransistor 343) and a fourth switch (N-type MOS transistors 344) whichexecute supply or blocking of supply of the ground VSS to the firstcurrent route and the second current route; and a comparative operationcontrol unit (comparative operation control circuit 340) for controllingsupply or blocking of the first through fourth switches.

Thus, supply or blocking timing of any of the first through fourthswitches is controlled, whereby the potentials of the first currentroute and the second current route before start of comparison can alsobe controlled.

As a result thereof, the potentials of the first current route and thesecond current route before start of comparison are controlled, wherebycomparison error along with different properties relating to the gatevoltages of on resistances of the first and second MOS transistors whichreceive input signals VIP and VIM can be controlled. Note that it goeswithout saying that comparison error can be controlled by controllingany of the first through fourth switches.

Further, the latch circuit portion includes the P-type MOS transistorlatch circuit 320 made up of a P-type MOS transistor (P-type MOStransistor 321) making up the first current route, and a P-type MOStransistor (P-type MOS transistor 322) making up the second currentroute. Note that supply and blocking of supply of the high-potential VDDto the P-type MOS transistor latch unit 320 are executed by the firstswitch and the second switch. The N-type MOS transistors making up theN-type MOS transistor latch unit are not included in the first currentroute and the second current route, and accordingly, capacitanceparasitizing the first current route and the second current routedecreases, and the response speed of the comparison circuit 300 becomesfast.

Fifth Embodiment

FIG. 9 is a circuit diagram illustrating a comparison circuit 400according to the fifth embodiment. The comparison circuit 400 includes aP-type MOS transistor latch unit 420, an input unit 430, an N-type MOStransistor latch unit 425, a comparative operation control circuit 440,and a comparative operation setting circuit 450.

The comparative operation control circuit 440 includes N-type MOStransistors 443 and 444, and P-type MOS transistors 441 and 442.

The P-type MOS transistor 441 has a drain to be connected to the sourceof a P-type MOS transistor 431 of the input unit 430, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LM1.

The P-type MOS transistor 442 has a drain to be connected to the sourceof a P-type MOS transistor 432 of the input unit 430, a source to beconnected to the high-potential VDD power source 60, and a gate forreceiving a signal LP1.

The P-type MOS transistor 441 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 430 when the logicof the signal LM1 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 430 whenthe logic is “H”. The P-type MOS transistor 441 serves as a switch forconnecting or blocking between the input unit 430 and the high-potentialVDD power source 60.

The P-type MOS transistor 442 supplies the high-potential VDD from thehigh-potential VDD power source 60 to the input unit 430 when the logicof the signal LP1 is “L”, and blocks supply of the high-potential VDDfrom the high-potential VDD power source 60 to the input unit 430 whenthe logic is “H”. The P-type MOS transistor 442 serves as a switch forconnecting or blocking between the input unit 430 and the high-potentialVDD power source 60.

The N-type MOS transistor 443 has a source to be connected to a groundVSS power source 70, a drain to be connected to the drain of the P-typeMOS transistor 431 of the input unit 430, and a gate for receiving asignal LM0.

Note that the drain of the N-type MOS transistor 443, the drain of theP-type MOS transistor 431 of the input unit 430, the drain of the P-typeMOS transistor 421 of the P-type MOS transistor latch unit 420, and thedrain of the N-type MOS transistor 426 of the N-type MOS transistorlatch unit 425 are all connected to the node A within the comparisoncircuit 400. The comparison circuit 400 outputs an output signal OM fromthe node A.

The N-type MOS transistor 443 supplies the ground VSS from the groundVSS power source 70 to the input unit 430 when the logic of the signalLM0 is “H”, and blocks supply of the ground VSS from the ground VSSpower source 70 to the input unit 430 when the logic is “L”. The N-typeMOS transistor 443 serves as a switch for connecting or blocking betweenthe input unit 430 and the ground VSS power source 70.

The N-type MOS transistor 444 has a source to be connected to the groundVSS power source 70, a drain to be connected to the drain of the P-typeMOS transistor 432 of the input unit 430, and a gate for receiving asignal LP0.

Note that the drain of the N-type MOS transistor 444, the drain of theN-type MOS transistor 432 of the input unit 430, the drain of the P-typeMOS transistor 422 of the P-type MOS transistor latch unit 420, and thedrain of the N-type MOS transistor 427 of the N-type MOS transistorlatch unit 425 are all connected to the node B within the comparisoncircuit 400. The comparison circuit 400 outputs an output signal OP fromthe node B.

The N-type MOS transistor 444 supplies the ground VSS from the groundVSS power source 70 to the input unit 430 when the logic of the signalLP0 is “H”, and blocks supply of the ground VSS from the ground VSSpower source 70 to the input unit 430 when the logic is “L”. The N-typeMOS transistor 444 serves as a switch for connecting or blocking betweenthe input unit 430 and the ground VSS power source 70.

The input unit 430 includes the P-type MOS transistors 431 and 432. TheP-type MOS transistor 431 has a drain to be connected to the drain ofthe P-type MOS transistor 421 of the P-type MOS transistor latch unit420, a source to be connected to the drain of the N-type MOS transistor443 of the comparative operation control circuit 440, and a gate forreceiving an input signal VIP. The on resistance value of the P-type MOStransistor 431 varies according to the potential of the input signalVIP.

The P-type MOS transistor 432 has a drain to be connected to the drainof the P-type MOS transistor 422 of the P-type MOS transistor latch unit420, a source to be connected to the drain of the N-type MOS transistor444 of the comparative operation control circuit 440, and a gate forreceiving a signal VIM. The on resistance value of the N-type MOStransistor 432 varies according to the potential of the signal VIM.

The P-type MOS transistor latch unit 420 includes the P-type MOStransistors 421 and 422. The N-type MOS transistor latch unit 425includes the N-type MOS transistors 426 and 427.

The P-type MOS transistor 421 has a drain to be connected to the drainof the N-type MOS transistor 426, a source to be connected to thehigh-potential VDD power source 60, and a gate to be connected to thedrain of the P-type MOS transistor 422.

The P-type MOS transistor 422 has a drain to be connected to the drainof the N-type MOS transistor 427, a source to be connected to thehigh-potential VDD power source 60, and a gate to be connected to thedrain of the P-type MOS transistor 421.

The N-type MOS transistor 426 has a drain to be connected to the drainof the P-type MOS transistor 421, a gate to be connected to the drain ofthe N-type MOS transistor 427, and a source to be connected to theground VSS 70.

The N-type MOS transistor 427 has a drain to be connected to the drainof the P-type MOS transistor 422, a gate to be connected to the drain ofthe N-type MOS transistor 426, and a source to be connected to theground VSS 70.

The output signals OM and OP are output from the N-type MOS transistorlatch unit 425. The output signal OM is connected to the node A betweenthe drain of the N-type MOS transistor 426 and the drain of the P-typeMOS transistor 421. The output signal OP is connected to the node Bbetween the drain of the N-type MOS transistor 427 and the drain of theP-type MOS transistor 422.

The gate of the N-type MOS transistor 426 of the N-type MOS transistorlatch unit 425 and the drain of the N-type MOS transistor 427 areconnected to the node B, and the gate of the N-type MOS transistor 427and the drain of the N-type MOS transistor 426 are connected to the nodeA. That is to say, the N-type MOS transistor 426 and the N-type MOStransistor 427 are connected crosswise to the nodes A and B, andaccordingly, the N-type MOS transistor 426 and the N-type MOS transistor427 amplify potential difference between the nodes A and B.

Thus, the N-type MOS transistor 426 of the N-type MOS transistor latchunit 425, the P-type MOS transistor 431, and the P-type MOS transistor441 are serially connected between the high-potential VDD power source60 and the ground VSS 70, and make up a first current route includingthe node A. The N-type MOS transistor 427 of the N-type MOS transistorlatch unit 425, the P-type MOS transistor 432, and the P-type MOStransistor 442 are serially connected between the high-potential VDDpower source 60 and the ground VSS 70, and make up a second currentroute including the node B.

Therefore, when the signals LM0, LM1, LP0, and LP1 are “H”, supply ofthe high-potential VDD to the input unit 430 is blocked by the P-typeMOS transistors 441 and 442, and the ground VSS is supplied to the inputunit 430, the P-type MOS transistor latch unit 420, and the N-type MOStransistor latch unit 425 by the N-type MOS transistors 443 and 444.

As a result thereof, potential difference between the nodes A and B is0, or almost eliminated.

On the other hand, when the signals LM0, LM1, LP0, and LP1 are “L”, thehigh-potential VDD is supplied to the input unit 430 by the P-type MOStransistors 441 and 442, and the supply of the ground VSS to the P-typeMOS transistor latch unit 420, the N-type MOS transistor latch unit 425,and the input unit 430 is blocked by the N-type MOS transistors 443 and444.

In the above case, the on resistance of the P-type MOS transistor 431varies according to the potential of the input signal VIP, andaccordingly, upon the potential of the input signal VIP decreasing, thepotential of the node A increases, and the on resistance of the N-typeMOS transistor 432 varies according to the potential of the input signalVIM, and accordingly, upon the potential of the input signal VIMincreasing contrary to the input signal VIP, the potential of the node Bdecreases.

On the other hand, conversely, upon the potential of the input signalVIP increasing, the potential of the node A decreases, and upon thepotential of the input signal VIM decreasing, the potential of the nodeB increases.

The comparative operation setting circuit 450 is made up of a circuitsimilar to the comparative operation setting circuit 50 according to thefirst embodiment for driving the signals LM0 and LP0, and a circuitsimilar to the comparative operation setting circuit 50 for driving thesignals LM1 and LP1. The circuits similar to the comparative operationsetting circuit 50 include a delay circuit 451, and a logic circuit 456for controlling the delay circuit 451. The delay circuit 451 and thelogic circuit 456 are the same circuits as the delay circuit 51 and thelogic circuit 56. Now, it goes without saying that delay between thesignals LM0 and LP0, and delay between the signals LM1 and LP1 may beset to the same delay amount, or may be set separately, and further, ofthe above delays, any one alone may be set.

Thus, the comparison circuit 400 according to the fifth embodimentincludes an input unit (430) made up of a first MOS transistor (P-typeMOS transistor 431) for receiving a first signal (signal VIP) at thegate electrode, and a second MOS transistor (P-type MOS transistor 432)for receiving a second signal (signal VIM) at the gate electrode; alatch circuit (P-type MOS transistor latch circuit 420, N-type MOStransistor latch circuit 425) for amplifying potential differencebetween a first current route where an electric current is controlled bythe first MOS transistor according to the voltage of the first signal,and a second current route where an electric current is controlled bythe second MOS transistor according to the voltage of the second signal;a comparative operation setting unit (comparative operation settingcircuit 450) including a first switch (P-type MOS transistor 441) forexecuting supply or blocking of supply of the high-potential VDD to thedrain of the first MOS transistor by a third current route differentfrom the first current route, a second switch (P-type MOS transistor442) for executing supply or blocking of supply of the high-potentialVDD to the drain of the second MOS transistor by a fourth current routedifferent from the second current route, and a third switch (N-type MOStransistor 443) and a fourth switch (N-type MOS transistor 444) whichexecute supply or blocking of supply of the ground VSS to the firstcurrent route and the second current route; and a comparative operationcontrol unit (comparative operation control circuit 440) for controllingsupply or blocking of the first through fourth switches.

Thus, supply or blocking timing of any of the first through fourthswitches is controlled, whereby the potentials of the first currentroute and the second current route before start of comparison can alsobe controlled.

As a result thereof, the potentials of the first current route and thesecond current route before start of comparison are controlled, wherebycomparison error along with different properties relating to the gatevoltages of on resistances of the first and second MOS transistors whichreceive input signals VIP and VIM can be controlled. Note that it goeswithout saying that comparison error can be controlled by controllingany of the first through fourth switches.

Further, the latch circuit includes the N-type MOS transistor latch unit425 made up of an N-type MOS transistor (N-type MOS transistor latchcircuit 426) making up the first current route, and an N-type MOStransistor (N-type MOS transistor 427) making up the second currentroute. Note that supply and blocking of supply of the ground VSS to theN-type MOS transistor latch unit 425 is executed by the first and secondswitches. The P-type MOS transistors making up the P-type MOS transistorlatch unit are not included in the first current route and the secondcurrent route, and accordingly, capacitance parasitizing the firstcurrent route and the second current route decreases, and the responsespeed of the comparison circuit 400 becomes fast.

Sixth Embodiment

FIG. 10 illustrates an ADC (Analog Digital Converter) 500 according tothe sixth embodiment. The ADC 500 according to the sixth embodimentincludes the latch unit 20, input unit 30, and the comparative operationcontrol circuit 40, of the comparison circuit 10 according to the firstembodiment, and further includes a delay circuit 520, a successivecomparative operation control circuit 530, and a sample-hold circuit540.

The ADC 500 according to the sixth embodiment is a successivecomparative type analog-to-digital conversion circuit using the latchunit 20, input unit 30, and comparative operation control circuit 40,described in the first embodiment.

Accordingly, detailed description will be omitted regarding the latchunit 20, input unit 30, and comparative operation control circuit 40, ofthe ADC 500 according to the sixth embodiment.

The input unit 30 receives a complementary input signal Vi, andgenerates an inversion complementary signal having inversion logic. Thelatch unit 20 receives and latches the inversion complementary signalthereof. The comparison control circuit 40 receives the signal from thesuccessive comparative operation control circuit 530, and connects ordisconnects the input unit 30 and the latch unit 20 to or from thehigh-potential power source Vcc according to the logic thereof. Thelatch unit 20 and the input unit 30 make up a comparator for comparingthe potential of a signal Vi+ and the potential of a signal Vi− whichmake up the complementary input signal Vi.

Note that though the latch unit 20, the input unit 30, and thecomparative operation control circuit 40 operate according to a signalCNTL531 output from the successive comparative operation control circuit530, details regarding the operation thereof will be described withreference to FIG. 12.

The delay circuit 520 is configured of inverters 521, 522, 526, and 527,inverters 523 and 525 using variable signal delay time, and an inverter524 which receives a signal CLK at the input terminal.

The inverter 524 receivers the signal CLK, and outputs an output signalA to a switch 43 a included in the comparative operation control circuit40 to connect or disconnect the ground potential and the input unit 30according to the logic of the output signal A.

The inverters 523 and 525 receive the output signal A of the inverter524 at the input terminals thereof. The inverter 522 receives the outputsignal of the inverter 523 at the input terminal thereof. The inverter521 receives the output signal of the inverter 522 at the input terminalthereof. The inverter 522 outputs a signal Axp to one switch 41 a of thecomparative operation control circuit 40. Note that the one switch 41 aconnects or disconnects the input unit 30 and the latch unit 20 from thehigh-potential power source according to the logic of the signal Axp.With the inverter 523, signal delay time from the signal having beenreceived at the input terminal until the output signal is output variesaccording to the signal CNTL531 made up of multiple digital signals.Examples of the inverter 523 will be shown in FIGS. 11A and 11B.

The inverter 526 receives the output signal of the inverter 525 at theinput terminal thereof. The inverter 527 receives the output signal ofthe inverter 526 at the input terminal thereof. The inverter 527 outputsa signal Axm to the other switch 42 a of the comparative operationcontrol circuit 40. Note that the other switch 42 a connects ordisconnects the input unit 30 and the latch unit 20 from thehigh-potential power source according to the logic of the signal Axm.With the inverter 525, signal delay time from the signal having beenreceived at the input terminal until the output signal is output variesaccording to the signal CNTL531 made up of multiple digital signals.Examples of the inverter 525 will be shown in FIGS. 11A and 11B.

Accordingly, with the binary numeral represented by the signal CNTL531,time difference can be set to between the output period of the signalAxm and the output period of the signal Axp.

The sample-hold circuit 540 includes a switch 541 for connecting ordisconnecting an input terminal which receives one signal Vi+ making upthe input signal Vin thereof, and a signal line to be connected to theinput unit 30 according to the logic of the signal CLK, and acapacitance 542 to be connected to this signal line. Also, thesample-hold circuit 540 includes a switch 543 for connecting ordisconnecting an input terminal which receives the other signal Vi−making up the input signal Vin thereof, and a signal line to beconnected to the input unit 30 according to the logic of a controlsignal CN532 from the successive comparative operation control circuit530, and a capacitance 544 to be connected to this signal line.

Accordingly, the sample-hold circuit 540 is a circuit for receiving thecomplementary input signal Vin, and sampling the voltage of one signalVi+ making up the input signal Vin thereof, and the voltage of thecomplementary signal Vi− thereof.

The successive comparative operation control circuit 530 receivessignals Vo+ and Vo− from the latch unit 20, and the signal CLK, outputsthe signal CNTL531 to the delay circuit 520, and outputs the controlsignal CN532 to the sample-hold circuit 540. Note that the operation ofthe successive comparative operation control circuit 530, and the signalCNTL531 will be described in detail with reference to FIGS. 11 through16. Also, the control signal CN532 will be described in detail withreference to FIG. 15.

FIGS. 11A and 11B illustrate examples of the inverters 523 and 525 ofthe delay circuit 520. FIG. 11A illustrates an inverter 528 which setsdelay time from input of an input signal until output of an outputsignal to be variable using multiple combination circuits made up of aresistor, and a switch for connecting or disconnecting the resistorthereof to or from the ground line. The inverter 528 is configured of ncombination circuits from a combination circuit 528 r 1 to a combinationcircuit 528 rn (n is a positive integer) made up of a P-type transistor528 a, an N-type transistor 528 b, a resistor, and a switch, an NOR 528c, an input terminal 528 d, and an output terminal 528 e.

The P-type transistor 528 a is a MOS transistor wherein the sourcethereof is connected to the high-potential power source Vcc, the drainthereof is connected to the output terminal 528 e, and the gate thereofis connected to the input terminal 528 d.

The P-type transistor 528 b is a MOS transistor wherein the sourcethereof is connected to the n combination circuits from the combinationcircuit 528 r 1 to the combination circuit 528 rn, the drain thereof isconnected to the output terminal 528 e, and the gate thereof isconnected to the input terminal 528 d.

The n combination circuits from the combination circuit 528 r 1 to thecombination circuit 528 rn are connected between the source of theN-type transistor 528 b and the ground Vss in parallel. Each of thecombination circuits is configured by a resistor, a first switch, and asecond switch being connected serially, and one end of the resistor isconnected to the source of the N-type transistor 528 b, and the otherend of the second switch is connected to the ground Vss.

If we say that the resistance value of the resistor of the combinationcircuit 528 r 1 is 1, the resistance value of the resistor of thecombination circuit 528 rn is the n'th power of 2. The signal CNTL531 ismultiple signals representing n+2 binary codes, and signals representingthe first bit to the n'th bit of the binary numeral that the signalCNTL531 represents are connected to the first switches from thecombination circuit 528 r 1 to the combination circuit 528 rn. The firstswitches from the combination circuit 528 r 1 to the combination circuit528 rn are connected when the signals of the signal CNTL531 are “1”, andare disconnected when the signals of the signal CNTL531 have an inverselogic.

The signal corresponding to the n+1'th bit of the binary numeral thatthe signal CNTL531 represents is commonly connected to all the secondswitches from the combination circuit 528 r 1 to the combination circuit528 rn. If we say that the second switch of the inverter 528 operatingas the inverter 523 operates to be connected when the logic of thesignal thereof is “1”, and operates to be disconnected when the logic is“0”, the second switch of the inverter 528 operating as the inverter 525operates to be connected when the logic of the signal thereof is “0”,and operates to be disconnected when the logic is “1”.

Also, the signal corresponding to the n+1'th bit of the binary numeralthat the signal CNTL531 represents is connected to one input terminal ofthe NOR 528 c. The NOR 528 c inverts the logic of the signal thereof,and outputs the output signal thereof to the switch 528 f. The switch528 f connects or disconnects the source of the N-type transistor 528 bto or from the ground Vss according to the logic of the output signalfrom the NOR 528 c.

The signal corresponding to the n+2'th bit of the binary numeral thatthe signal CNTL531 represents is connected to the other input terminalof the NOR 528 c. The signal corresponding to the n+2'th bit of thesignal CNTL531 is used when both of the signal Axp and the signal Axmare output without delay from the signal A.

Thus, with the inverter 528, operation that makes delay time from inputof an input signal to output of an output signal variable according tothe signal representing the n+1'th bit of the signal CNTL531, andoperation that makes the delay time constant are switched. With theinverter 528, in the case of making the delay time variable, themagnitude of the delay time increases according to the magnitude of thebinary numeral represented by from bit 1 to bit n of the signal CNTL531.This is because the greater the magnitude of the binary numeralrepresented by from bit 1 to bit n of the CNTL531 is, the smaller theelectric current from the ground Vss flowing to the source of the N-typetransistor 528 b is.

FIG. 11B illustrates an inverter 529 which makes the delay time frominput of an input signal to output of an output signal variable usingmultiple combination circuits made up of a resistor, and a switch forconnecting or disconnecting the capacitance thereof to or from a groundline. The inverter 529 is configured of a P-type transistor 529 a, anN-type transistor 529 b, n combination circuits from a combinationcircuit 529 c 1 to a combination circuit 529 cn (n is a positiveinteger) which are made up of capacitance and a switch, an inputterminal 529 d, a NOR 529 f, and an output terminal 529 e.

The P-type transistor 529 a is a MOS transistor wherein the sourcethereof is connected to the high-potential power source Vcc, the drainthereof is connected to the output terminal 529 e, and the gate thereofis connected to the input terminal 529 d.

The N-type transistor 529 b is a MOS transistor wherein the sourcethereof is connected to the ground Vss, the drain thereof is connectedto the output terminal 529 e, and the gate thereof is connected to theinput terminal 528 d.

The n combination circuits from the combination circuit 529 c 1 to thecombination circuit 529 cn (n is a positive integer) are connectedbetween the output terminal 529 e and the ground Vss in parallel. Eachof the combination circuits is configured by capacitance, a firstswitch, and a second switch being serially connected, and one end of thecapacitance is connected to the output terminal 529 e, and the other endof the second switch is connected to the ground Vss.

If we say that the capacitance value of the capacitance of thecombination circuit 529 c 1 is 1, the capacitance value of thecapacitance of the combination circuit 529 cn is the n'th power of 2.The signal CNTL531 is multiple signals representing binary numeral ofn+2 bits, and signals representing the first bit to the n'th bit of thebinary numeral that the signal CNTL531 represents are connected to thefirst switches from the combination circuit 529 c 1 to the combinationcircuit 529 cn. The first switches from the combination circuit 529 c 1to the combination circuit 529 cn are connected when the signals of thesignal CNTL531 are “1”, and are disconnected when the signals of thesignal CNTL531 have an inverse logic.

The signal corresponding to the n+1'th bit of the binary numeral thatthe signal CNTL531 represents is connected to one terminal of the NOR529 f, and the output of the NOR 529 f is commonly connected to all thesecond switches from the combination circuit 529 c 1 to the combinationcircuit 529 cn. If we say that the second switch of the inverter 529operating as the inverter 523 operates to be connected when the logic ofthe signal thereof is “1”, and operates to be disconnected when thelogic is “0”, the second switch of the inverter 529 operating as theinverter 525 operates to be connected when the logic of the signalthereof is “0”, and operates to be disconnected when the logic is “1”.

The signal corresponding to the n+2'th bit of the binary numeral thatthe signal CNTL531 represents is connected to the other input terminalof the NOR 529 f. The signal corresponding to the n+2'th bit of thesignal CNTL531 is used when both of the signal Axp and the signal Axmare output without delay from the signal A.

Thus, with the inverter 529, operation that makes delay time from inputof an input signal to output of an output signal variable according tothe signal representing the most significant bit of the signal CNTL531,and operation that makes the delay time constant are switched. With theinverter 529, in the case of making the delay time variable, themagnitude of the delay time increases according to the magnitude of thebinary numeral represented by bit 1 to bit n of the signal CNTL531. Thisis because the greater the magnitude of the binary numeral representedby bit 1 to bit n of the CNTL531 is, the greater the capacitance valueof the capacitance to be connected to the output terminal 529 e of theN-type transistor is.

FIGS. 12A and 12B are diagrams for describing the operation according tothe sixth embodiment regarding the latch unit 20, input unit 30,comparative operation control circuit 40, and delay circuit 520.

With the first embodiment, the latch unit 20, input unit 30, comparativeoperation control circuit 40, and delay circuit 520 make up a comparatorwhich operates so as to compare the magnitude between the potential ofthe signal Vi+ and the potential of the signal Vi− making up thecomplementary input signal Vi input to the input unit 30.

However, with the sixth embodiment, operation for comparing themagnitude of the potential of an input signal with the first embodimentis a basic operation, and further, this basic operation is repeatedwhile shifting the leading or trailing-edge period of the signals Axpand Axm, and accordingly, the latch unit 20, input unit 30, comparativeoperation control circuit 40, and delay circuit 520 operate as a circuitfor detecting the complementary input signal input to the input unit 30,i.e., the potential difference of the complementary signal made up ofthe signal Vi+ and the signal Vi−.

FIG. 12A is a diagram illustrating the potential waveforms of theprincipal signals of the latch unit 20, input unit 30, comparativeoperation control circuit 40, and delay circuit 520, corresponding tothe operation for shifting the trailing-edge period of the signals Axpand Axm.

The successive comparative operation control circuit 530 outputs thesignal CNTL5312 having a logic representing time difference between theleading or trailing-edge periods of the signals Axp and Axm duringpoint-in-time T1 through point-in-time T5 to cause the latch unit 20,input unit 30, comparative operation control circuit 40, and delaycircuit 520 to execute one-time basic operation. Such as described inFIGS. 11A and 11B, whether the signal Axm or signal Axp is delayed isdetermined according to the logic of the most significant bit of thesignal CNTL531. FIG. 12A illustrates an example wherein the signal Axmis delayed.

Thus, the delay circuit 520 delays and outputs the signal A and thesignal A synchronized with the signal CLK, and also outputs the signalAxm according to the logic of the signal CNTL531. The leading edge ofthe signal A is point-in-time T2, and the trailing edges thereof arepoint-in-time T1 and point-in-time T5. The leading edges of the signalAxp are point-in-time T1 and point-in-time T5, and the trailing edgesthereof are point-in-time T3. Also, the leading edge of the leadingedges of the signal Axm are point-in-time T1 and point-in-time T5, andthe trailing edge thereof is point-in-time T3. Here, time differencebetween the point-in-time T2 and point-in-time T3 indicatestrailing-edge time difference between the signals Axp and Axm.

Note that FIG. 12A illustrates the operation of each circuit for onecycle of the signal CLK, and illustrates the operation of each circuitin the case that the trailing-edge time difference between the signalsAxp and Axm represented by the signal CNTL is 0.21 ns. However, beforethe point-in-time T1, the successive comparative operation controlcircuit 530 increases the trailing-edge time difference between thesignals Axp and Axm represented by the logic of the signal CNTL inincrements of 0.01 ns for each two cycle.

The signal Vi to be input to the input unit 30 is a complementarysignal, and in FIG. 12A, the difference between the signal Vi+ andsignal Vi− making up the complementary input signal Vi is kept in 100mv.

At the point-in-time T1, upon the logic of the signal A turning to “L”,and the logics of the signals Axp and Axm turning to “H”, the switch ofthe comparative operation control circuit 40 connects the latch unit 20and the high-potential power source Vcc, and connects the input unit 30and the ground Vss, and accordingly, the logics of the signals Vp and Vmlatched by the latch unit 20 both turn to “H”.

Subsequently, at the point-in-time T2, upon the logic of the signal Aturning to “H”, and the logic of the signal Axp turning to “L”, thelogic of the signal Vm begins to operate toward “L”. On the other hand,at the point-in-time T3, the logic of the signal Axm also turns to “L”,and the logic of the signal Vp also begins to operate toward “L”. Here,the potential of the signal Vi− is lower than the potential of thesignal Vi+, and accordingly, speed wherein the logic of the signal Vpproceeds to “L” is faster than speed wherein the logic of the signal Vmproceeds to “L”. However, point-in-time when the logic of the signal Vpbegins to proceed to “L” is T3 that is slower than T2, and accordingly,the potential of the signal Vm first exceeds the operation threshold ofthe latch unit 20. As a result thereof, at point-in-time T4, accordingto the operation of the latch unit 20, the logic of the signal Vm turnsto “L”, and the logic of the signal Vp turns to “H”. As a resultthereof, the logic of an output signal Vo changes at the point-in-timeT3.

Subsequently, at point-in-time T5, upon the logic of the signal Aturning to “L”, and the logics of the signals Axp and Axm turning to “H”again, the latch unit 20, input unit 30, comparative operation controlcircuit 40, and delay circuit 520 return to the state at thepoint-in-time again.

FIG. 12B is a diagram representing change in the signal Vm whenrepeating the operation from the point-in-time T1 to point-in-time T5while changing trailing-edge time difference td between the signals Axpand Axm from 0.16 ns to 0.25 ns.

In FIG. 12B, the vertical axis represents potentials, and the horizontalaxis indicates elapse of time. Thin solid lines indicate thetrailing-edge point-in-time of the signal CLK. Also, heavy solid linesindicate change in the potential of the signal Vm.

For example, when the time difference td of the trailing edges of thesignals Axp and Axm represented by the logic of the signal CNTL is 0.2ns, this indicates that the potential of the signal Vm has changedbetween 3 V and 1.8 V.

Also, when the time difference td is 0.21 ns, this indicates that thepotential of the signal Vm has changed between 3 V and 0 V.

In the event that the time difference td is equal to or smaller than0.20 ns, this indicates that the logic of the signal Vm latched by thelatch unit 20 is “H” after comparative operation. Also, upon the timedifference td reaching equal to or greater than 0.21 ns, this indicatesthat the logic of the signal Vm latched by the latch unit 20 is “L”after comparative operation. That is to say, the logic of the signal Vmor signal Vp after latch changes with time difference td=0.21 ns as aborder.

Thus, in the case that there is potential difference between the signalVi− and signal Vi+, upon observing change in the logic of the signal Vmor signal Vp latched by the latch unit 20 while changing the timedifference td of the trailing edges of the signals Axp and Axmrepresented by the logic of the signal CNTL, the time difference td tocause change in the logic of the signal Vm or signal Vp as to the abovetime difference td after comparative operation can be detected.

Therefore, it can be found that relationship of the potential differencebetween the signal Vi+ and signal Vi+ making up the input signal Vi, andthe time difference td can be obtained beforehand.

Further, relationship of the logic of the signal CNTL, and the timedifference td of the trailing edges of the signals Axp and Axm isuniquely determined such as described in FIGS. 11A and 11B, andaccordingly, relationship of the potential difference between the signalVi− and signal Vi+, and the logic of the signal CNTL is uniquelydetermined.

FIG. 13 is a diagram representing relationship of the logic of thesignal CNTL, the time difference td of the trailing edges of the signalAxp and signal Axm, the potential difference between the signal Vi− andsignal Vi+ making up the input signal Vi.

In FIG. 13, the horizontal axis on the lower side represents the timedifference td, the vertical axis represents the potential differencebetween the signal Vi− and signal Vi+ making up the input signal Vi, andthe horizontal axis on the upper side indicates the binary numeral thatthe signal CNTL 531 corresponding to the time difference td represents.

According to FIG. 13, the relationship of the potential differencebetween the signal Vi− and signal Vi+, and the time difference td of thetrailing edges of the signals Axp and Axm used for changing the logic ofthe signal Vm or Vp indicates monotone increase. In more detail, forexample, if we represent this relationship such as (potentialdifference, td), relationship of (0.01, 0.15), (0.05, 0.17), (0.1, 0.2),(0.2, 0.23), (0.3, 0.27), (0.4, 0.3), (0.5, 0.33), (0.6, 0.37), (0.7,0.39), and (0.8, 0.41) is shown. Note that relationship of “the timedifference td” and “the potential difference between the signal Vi− andsignal Vi+” can be obtained by circuit simulation based on the circuitdiagram of the latch unit 20, input unit 30, comparative operationcontrol circuit 40, and delay circuit 520 shown in FIG. 10.

The binary numerals shown in the upper portion of the drawing aredisposed in the positions corresponding to “the time difference td ofthe trailing edges of the signals Axp and Axm” obtained when the signalCNTL531 representing the binary numerals thereof is input. The upperstage of the binary numerals shown in the upper portion of the drawingindicates binary numerals used for delaying only the trailing-edgeperiod of the signal Axm with the trailing-edge period of the signal Axpbeing fixed as to the trailing-edge period of the signal A. On the otherhand, the binary numerals on the lower stage indicate binary numeralsused for delaying only the trailing-edge period of the signal Axp withthe trailing-edge period of the signal Axm being fixed as to thetrailing-edge period of the signal A.

If we describe the correspondence as (binary, tpd), for example, thisyields correspondences of (00000, 0.12 ns), (00001, 0.12 ns), (10000,0.14 ns), (10001, 0.14 ns), (01000, 0.16 ns), (01001, 0.16 ns), (11000,0.18 ns), (11001, 0.18 ns), (00100, 0.20 ns), (00101, 0.20 ns), (10100,0.22 ns), (10101, 0.22 ns), (01100, 0.24 ns), (01101, 0.24 ns), (11100,0.26 ns), (11101, 0.26 ns), (00010, 0.28 ns), (00011, 0.28 ns), (10010,0.30 ns), (10011, 0.30 ns), (01010, 0.32 ns), (01011, 0.32 ns), (11010,0.34 ns), (11011, 0.34 ns), (00110, 0.36 ns), (00111, 0.36 ns), (10110,0.38 ns), (10111, 0.38 ns), (01110, 0.40 ns), (01111, 0.40 ns), (11110,0.42 ns), and (11111, 0.42 ns).

Note that the above correspondences can be obtained by executing circuitsimulation regarding the inverters 528 and 529 shown in FIGS. 11A and11B.

FIG. 14 illustrates an operation waveform when detecting the potentialdifference between the complementary signal made up of the signal Vi+and signal Vi− by the ADC circuit 500 according to the sixth embodiment.

The signal CLK is a clock signal which repeats logic “H” and “L” in aconstant cycle. The control signal CN532 is a control signal to beoutput from the successive comparative operation control circuit 530.When the logic of the control signal CN532 of the sample-hold circuit540 is “H”, i.e., during from the point-in-time T1 to the point-in-timeT3, when the switches 541 and 543 connect the input terminal and thecapacitances 542 and 544, sample the potentials of the signal Vi+ andsignal Vi−, and the logic of the control signal CNT532 is “L”, i.e.,during from the point-in-time T3 to the point-in-time T8, and duringfrom the point-in-time T10 to the point-in-time T12, the potentials ofthe signal Vi+ and signal Vi− are held.

The signal A is a logic inversion signal of the signal CLK synchronizedwith the signal CLK, to be output from the inverter 524 which receivedthe signal CLK. The signal Axp is output from the inverter 524 whichreceived the signal CLK to the switch 41 a of the comparative controlcircuit 40 via the inverters 523, 522, and 521. The signal Axm is outputfrom the inverter 524 which received the signal CLK to the switch 42 aof the comparative control circuit 40 via the inverters 525, 526, and527.

If we say that signals making up the complementary signal Vo are onesignal Vo+ and the other signal Vo−, these signals have the same voltagewhen the signal A falls. On the other hand, when the signal A rises, thelatch unit 20 latches the logic of the signal Vm and the logic of thesignal Vp, and outputs these to the successive comparative operationcontrol circuit 530 as signals Vo+ and Vo−.

The signal CNTL531 is output from the successive comparative operationcontrol circuit 530 to the inverter 523 or inverter 525 in sync with thesignal CLK (i.e., from the input signal Vi having been sampled and helduntil point-in-time T12 from point-in-time T3 for each point-in-time).The signal CNTL531 represents a binary numeral determining td=|Axp−Axm|.Accordingly, the latch unit 20, input unit 30, comparative operationcontrol circuit 40, and delay circuit 520 executes comparison betweenthe potential of the signal Vi+ and the potential of the signal Vi−using td=|Axp−Axm| controlled by the successive comparative operationcontrol circuit 530. Note that description will be made regarding amethod for detecting and digitizing difference between the potential ofthe signal Vi+ and the potential of the signal Vi− according to thecontrol of the successive comparative operation control circuit 530, andcontrol thereof, with reference to the flowchart in FIG. 15.

After the operations of a series of the latch unit 20, input unit 30,comparative operation control circuit 40, and delay circuit 520, adigital signal Dout representing the difference between the potential ofthe signal Vi+ and the potential of the signal Vi− detected by thesuccessive comparative operation control circuit 530 is output by thesuccessive comparative operation control circuit 530 at thepoint-in-time T2 and point-in-time T9.

FIG. 15 is a flowchart for describing the control of td=|Axp−Axm|executed by the successive comparative operation control circuit 530,and a method for detecting the difference between the potential of thesignal Vi+ and the potential of the signal Vi− executed by the controlthereof.

s600: The successive comparative operation control circuit 530 outputs acontrol signal 532 to the sample-hold circuit 540, and samples and holdsthe input signal Vi. Next, the flow proceeds to s605.

s605: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, and sets td=|Axp−Axm|=0.That is to say, delays of the signals Axp and Axm from the signal A are0 ns. Next, the flow proceeds to s610.

s610: The delay circuit 520 executes signal output operation forreceiving the signal CLK, and outputting the signal A, signal Axp, andsignal Axm. Subsequently, the successive comparative operation controlcircuit 530 receives the signal Vo+ and signal Vo− output from the latchunit 20, and when the logic of the signal Vo+ is “H”, determines thatthe potential of the signal Vi+ is higher than the potential of thesignal Vi−, sets the most significant bit to “1”, and the flow proceedsto s615. On the other hand, when the logic of the signal Vo+ is “L”, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi+ is lower than the potential of the signalVi−, sets the most significant bit to “0”, and the flow proceeds tos685.

s615: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.3 ns, and sets td=|Axp−Axm|=0.3 ns. Next, the flowproceeds to s620.

s620: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.4 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the second bit to “1”, and the flow proceeds to s625. On theother hand, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is lower than thepotential +0.4 V of the signal Vi− when the logic of the signal Vo+ is“L”, sets the second bit to “0”, and the flow proceeds to s630.

s625: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.36 ns, and sets td=|Axp−Axm|=0.36 ns. Next, the flowproceeds to s635.

s635: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.6 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the third bit to “1”, and the flow proceeds to s645. On theother hand, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is lower than thepotential +0.6 V of the signal Vi− when the logic of the signal Vo+ is“L”, sets the third bit to “0”, and the flow proceeds to s650.

s645: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.38 ns, and sets td=|Axp−Axm|=0.38 ns. Next, the flowproceeds to s665.

s665: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.7 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the fourth bit to “1”, and outputs the digital signal Doutrepresenting a binary numeral (1111). On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi+ is lower than the potential +0.7 V of the signal Vi−when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, andoutputs the digital signal Dout representing a binary numeral (1110).

s650: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.34 ns, and sets td=|Axp−Axm|=0.34 ns. Next, the flowproceeds to s670.

s670: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.5 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the fourth bit to “1”, and outputs the digital signal Doutrepresenting a binary numeral (1101). On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi+ is lower than the potential +0.5 V of the signal Vi−when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, andoutputs the digital signal Dout representing a binary numeral (1100).

s630: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.22 ns, and sets td=|Axp−Axm|=0.22 ns. Next, the flowproceeds to s640.

s640: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.2 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the third bit to “1”, and the flow proceeds to s660. On theother hand, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is lower than thepotential +0.2 V of the signal Vi− when the logic of the signal Vo+ is“L”, sets the third bit to “0”, and the flow proceeds to s655.

s660: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.26 ns, and sets td=|Axp−Axm|=0.26 ns. Next, the flowproceeds to s680.

s680: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.3 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the fourth bit to “1”, and outputs the digital signal Doutrepresenting a binary numeral (1011). On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi+ is lower than the potential +0.3 V of the signal Vi−when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, andoutputs the digital signal Dout representing a binary numeral (1010).

s655: The successive comparative operation control circuit 530 outputsthe signal CNTL531 to the delay circuit 520, sets delay of the signalAxp from the signal A to 0 ns, sets delay of the signal Axm from thesignal A to 0.18 ns, and sets td=|Axp−Axm|=0.18 ns. Next, the flowproceeds to s675.

s675: The delay circuit 520 executes signal output operation similar tos610. Next, the successive comparative operation control circuit 530determines that the potential of the signal Vi+ is higher than thepotential +0.1 V of the signal Vi− when the logic of the signal Vo+ is“H”, sets the fourth bit to “1”, and outputs the digital signal Doutrepresenting a binary numeral (1001). On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi+ is lower than the potential +0.1 V of the signal Vi−when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, andoutputs the digital signal Dout representing a binary numeral (1000).

s685: The successive comparative operation control circuit 530 executesthe same operation as with s615, and sets delay of the signal Axp fromthe signal A to 0.3 ns. Next, the flow proceeds to s690.

s690: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s620, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.4 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the second bitto “0”, and the flow proceeds to s695. On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi− is lower than the potential +0.4 V of the signal Vi+when the logic of the signal Vo+ is “H”, sets the second bit to “1”, andthe flow proceeds to s700.

s695: The successive comparative operation control circuit 530 executesthe same operation as with s625, and sets delay of the signal Axp fromthe signal A to 0.36 ns. Next, the flow proceeds to s705.

s705: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s635, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.6 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the third bitto “0”, and the flow proceeds to s715. On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi− is lower than the potential +0.6 V of the signal Vi+when the logic of the signal Vo+ is “H”, sets the third bit to “1”, andthe flow proceeds to s720.

s715: The successive comparative operation control circuit 530 executesthe same operation as with s645, and sets delay of the signal Axm fromthe signal A to 0.38 ns. Next, the flow proceeds to s735.

s735: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s665, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.7 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bitto “0”, and outputs the digital signal Dout representing a binarynumeral (0000). On the other hand, the successive comparative operationcontrol circuit 530 determines that the potential of the signal Vi− islower than the potential +0.7 V of the signal Vi+ when the logic of thesignal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digitalsignal Dout representing a binary numeral (0001).

s720: The successive comparative operation control circuit 530 executesthe same operation as with s650, and sets delay of the signal Axp fromthe signal A to 0.34 ns. Next, the flow proceeds to s740.

s740: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s670, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.5 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bitto “0”, and outputs the digital signal Dout representing a binarynumeral (0010). On the other hand, the successive comparative operationcontrol circuit 530 determines that the potential of the signal Vi− islower than the potential +0.5 V of the signal Vi+ when the logic of thesignal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digitalsignal Dout representing a binary numeral (0011).

s700: The successive comparative operation control circuit 530 executesthe same operation as with s630, and sets delay of the signal Axp fromthe signal A to 0.22 ns. Next, the flow proceeds to s710.

s710: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s660, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.2 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the third bitto “0”, and the flow proceeds to s730. On the other hand, the successivecomparative operation control circuit 530 determines that the potentialof the signal Vi− is lower than the potential +0.2 V of the signal Vi+when the logic of the signal Vo+ is “H”, sets the third bit to “1”, andthe flow proceeds to s725.

s730: The successive comparative operation control circuit 530 executesthe same operation as with s660, and sets delay of the signal Axp fromthe signal A to 0.26 ns. Next, the flow proceeds to s750.

s750: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s680, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.3 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bitto “0”, and outputs the digital signal Dout representing a binarynumeral (0100). On the other hand, the successive comparative operationcontrol circuit 530 determines that the potential of the signal Vi− islower than the potential +0.3 V of the signal Vi+ when the logic of thesignal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digitalsignal Dout representing a binary numeral (0101).

s725: The successive comparative operation control circuit 530 executesthe same operation as with s675, and sets delay of the signal Axp fromthe signal A to 0.18 ns. Next, the flow proceeds to s745.

s745: The successive comparative operation control circuit 530 and thedelay circuit 520 execute the same operation as with s645, thesuccessive comparative operation control circuit 530 determines that thepotential of the signal Vi− is higher than the potential +0.1 V of thesignal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bitto “0”, and outputs the digital signal Dout representing a binarynumeral (0110). On the other hand, the successive comparative operationcontrol circuit 530 determines that the potential of the signal Vi− islower than the potential +0.1 V of the signal Vi+ when the logic of thesignal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digitalsignal Dout representing a binary numeral (0111).

FIGS. 16A and 16B are tables for describing a method for derivingrelationship between the binary numeral that the signal CNTL531represents, and the binary numeral that the digital signal Dout to beoutput by the analog-to-digital conversion by the ADC 500 represents inthe case that there is no linearity with correlation with differencebetween the potential of the signal Vi− and the potential of the signalVi+.

FIG. 16A is a table indicating the signal CNTL531, correlation with thedifference between the potential of the signal Vi− and the potential ofthe signal Vi+, and relationship between the signal CNTL531 and thebinary numeral that the digital signal Dout represents.

With the table shown in FIG. 16A, a first column indicates binarynumerals that the signal CNTL531 represents, a second column indicatestime difference td between the trailing edge of the signal Axp and thetrailing edge of the signal Axm, a third column indicates difference ΔVibetween the potential of the signal Vi− and the potential of the signalVi+, and a fourth column indicates binary numeral codes (binarynumerals) that the signal Dout represents, i.e., indicates resultsobtained by converting the analog value indicating the difference ΔVibetween the potential of the signal Vi− and the potential of the signalVi+ into a digital value by the ADC 500.

Therefore, such as the following, in the case that there is no linearitywith correlation between the binary numeral that the signal CNTL531represents, and the difference between the potential of the signal Vi−and the potential of the signal Vi+, analog-to-digital conversion isexecuted.

First, with regard to the inverter circuit shown in FIGS. 11A and 11B,according to circuit simulation, the td in the second column is obtainedwhile changing the binary numeral represented by the signal CNTL531 inthe first column from (11110) to (00000), and further from (00001) to(11111).

Next, circuit simulation is executed such as shown in FIGS. 12A and 12B,and when the input signal Vi is input so as to obtain the ΔVi in thethird column, tPD is obtained whereby the logics of the signals Vm andVp are inverted, and such as shown in FIG. 13, correlation data of thedifference between the potential of the signal Vi− and the potential ofthe signal Vi+, and the binary numeral represented by the signal CNTL531is obtained.

Next, such as the table in FIG. 16A, the signal CNTL531 corresponding toan equal interval point of the difference value between the potential ofthe signal Vi− and the potential of the signal Vi+ is obtained form thetable such as 0.1, 0.2, and so on through 0.8.

Next, in accordance with the flowchart in FIG. 15, analog-to-digitalconversion is executed using the signal CNTL531 corresponding to theequal interval point to obtain the digital value represented by thesignal Dout shown in the fourth column, i.e., binary numeral.

FIG. 16B is a table indicating correlation between the signal CNTL531,and the difference between the potential of the signal Vi− and thepotential of the signal Vi+, and relationship between the signal CNTL531and the binary numerals that the digital signal Dout represents.

With the table shown in FIG. 16B, a first column indicates thedifference ΔVi between the potential of the signal Vi− and the potentialof the signal Vi+, a second column indicates time difference td betweenthe trailing edge of the signal Axp and the trailing edge of the signalAxm, a third column indicates the binary numerals represented by thesignal CNTL531, and a fourth column indicates binary codes (binarynumerals) that the signal Dout represents, i.e., indicates resultsobtained by converting the analog value indicating the difference ΔVibetween the potential of the signal Vi− and the potential of the signalVi+ into a digital value by the ADC 500.

Therefore, such as the following, in the case that there is no linearitywith correlation between the binary numeral that the signal CNTL531represents, and the difference between the potential of the signal Vi−and the potential of the signal Vi+, analog-to-digital conversion isexecuted.

First, with regard to the inverter circuit shown in FIGS. 11A and 11B,according to the same method as with the description in FIG. 16A,correlation data between the difference between the potential of thesignal Vi− and the potential of the signal Vi+, and the binary numeralrepresented with the signal CNTL531 is obtained.

Next, in accordance with the flowchart in FIG. 15, three times ofadditions or deletions are changed using the signal CNTL531, and thelast signal CNTL531 is stored.

Next, the binary numeral of the signal Dout corresponding to the lastsignal CNTL531 is determined using the table in FIG. 16B.

Thus, as in FIG. 13, even in the event that correlation between thesignal CNTL531, and the difference between the potential of the signalVi− and the potential of the signal Vi+ has no proportionality relation,analog-to-digital conversion can be executed with the successivecomparative operation in FIG. 15.

Thus, the ADC 500 according to the sixth embodiment is a successivecomparative type analog-to-digital conversion device including acomparator including an input unit (input unit 30) for receiving acomplementary input signal to generate an inversion complementary signalhaving the inversion logic of the complementary input signal, and alatch unit (latch unit 20) for latching the inversion complementarysignal; a comparative operation control circuit 40 including a firstswitch (switch 41 a) for connecting or disconnecting the latch unit andthe input unit to or from a high-potential power source according to thelogic of a first signal (signal Axp or signal Axm), and a second switch(switch 42 a) for connecting or disconnecting the latch unit and theinput unit to or from the high-potential power source according to thelogic of a second signal (signal Axp or signal Axm); a delay circuit 520for outputting the first signal (signal Axp or signal Axm), and thesecond signal (signal Axp or signal Axm); and a successive operationcontrol circuit for outputting a control signal for controlling theperiod of logic change of the first signal for controlling disconnectionof the first switch (switch 41 a), and the period of logic change of thesecond signal for controlling disconnection of the second switch (switch42 a) based on the logic of a signal latched by the latch unit.

With a common successive comparative type analog-to-digital conversiondevice, in order to convert potential difference between the potentialof the signal Vi+ and the potential of the signal Vi− making up thecomplementary input signal Vi, the difference between the potential ofthe signal Vi+ and the potential of the signal Vi− has to be determinedaccording to a series of operation for comparing both potentials whilechanging the degree of potential increase/decrease as to one of thepotential of the sampled signal Vi+, and the potential of the signalVi−.

When executing increase/decrease in potential, a method has to be usedwherein electric charge secured at the time of sampling of a signal isstored. Therefore, with a common successive comparative typeanalog-to-digital conversion device, one electrode of capacitance isconnected to the node in which the electric charge secured at the timeof sampling of a signal is sealed, and voltage applied to the otherelectrode is changed, thereby changing the degree of increase/decreasein potential.

Thus, with a common successive comparative type analog-to-digitalconversion device, capacitance to be connected to the node in which theelectric charge secured at the time of sampling of a signal is sealedresults in increase in circuit area.

However, with the ADC 500 according to the sixth embodiment, the sameeffects as with a case where increase/decrease in potential is added toone of the potential of the signal Vi+ and the potential of the signalVi− with potential comparative operation are created by providingdifference between the disconnection period of the first switch to beconnected to the drain of an NMOS transistor which receives the signalVi+ of the input unit 30, and the disconnection period of the secondswitch to be connected to the drain of an NMOS transistor which receivesthe signal Vi− of the input unit 30.

Thus, capacitance to be connected to the node in which the electriccharge secured at the time of sampling of a signal is sealed can beeliminated from the ADC 500 according to the sixth embodiment.Accordingly, the circuit area of the ADC 500 according to the sixthembodiment can be reduced as compared to a common successive comparativetype analog-to-digital circuit.

The ADC 500 according to the sixth embodiment is further ananalog-to-digital conversion device including a circuit for sampling andholding one signal and the other signal of the complementary inputsignal.

The delay circuit 520 of the ADC 500 according to the sixth embodimentis an analog-to-digital conversion device including a first circuit foroutputting the first signal, and a second circuit for outputting thesecond signal, the first circuit and the second circuit include aninverter circuit, and the inverter circuit includes a logic inversioncircuit for inverting the logic of a signal to be input, and a circuitfor adding load capacitance to the output signal line of the logicinversion circuit according to the binary numeral represented by thecontrol signal.

The delay circuit 520 of the ADC 500 according to the sixth embodimentis an analog-to-digital conversion device including a first circuit foroutputting the first signal, and a second circuit for outputting thesecond signal, the first circuit and the second circuit include aninverter circuit, and the inverter circuit includes a logic inversioncircuit for inverting the logic of a signal to be input, and a circuitfor making resistance between the logic inversion circuit and a groundpower source line variable according to the binary numeral representedby the control signal.

Seventh Embodiment

With the sixth embodiment, the switches 41 a and 42 a of the comparativeoperation control circuit 40 have been single switches. However, theswitches 41 a and 42 a of an ADC 600 according to the seventh embodimentinclude multiple switches.

FIG. 17 illustrates the ADC 600 according to the seventh embodiment. TheADC 600 according to the seventh embodiment includes a latch unit 20, aninput unit 30, a comparative operation control circuit 640, a delaycircuit 620, a successive comparative operation control circuit 630, anda sample-hold circuit 540.

The latch unit 20, input unit 30, and sample-hold circuit 540 are thesame circuits described in the sixth embodiment, and descriptionregarding the configuration and operation thereof will be omitted.

The successive comparative operation control circuit 630 causes thelatch unit 20, input unit 30, comparative operation control circuit 640,and delay circuit 620 to execute operation shown in the flowchart inFIG. 15 for comparing the potential of the signal Vi+ and the potentialof the signal Vi− making up the complementary input signal Vi in syncwith the signal CLK. As described later, in order to execute selectionof one of signals Axp1 and Axp2, and selection of one of signals Axm1and Axm2, to be output from the delay circuit 620, the number of bits ofthe binary numeral represented by a signal CNTL631 that the successivecomparative operation control circuit 630 outputs is greater than thenumber of bits of the signal CNTL531 according to the sixth embodimentby one bit.

The delay circuit 620 includes inverters 621 a, 622 a, and 623 a whichoutput the signal Axp1, inverters 621 b, 622 b, and 623 b which outputthe signal Axp2, inverters 624 a, 625 a, and 626 a which output thesignal Axm1, and inverters 624 b, 625 b, and 626 b which output thesignal Axm2. Here, time difference between the logic trailing edge ofthe signal A and the trailing edge of the signal Axp1, and timedifference between the logic trailing edge of the signal A and thetrailing edge of the signal Axp2 are changed according to the binarynumeral represented by the signal CNTL631.

Also, time difference between the logic trailing edge of the signal Aand the trailing edge of the signal Axm1, and time difference betweenthe logic trailing edge of the signal A and the trailing edge of thesignal Axm2 are changed according to the binary numeral represented bythe signal CNTL631.

The comparative operation control circuit 640 includes switches 41 b and41 c for connecting or disconnecting the latch unit 20 and the inputunit 30 to or from the high-potential power source Vcc according to thelogics of the signals Axp1 and Axp2, switches 42 b and 42 c forconnecting or disconnecting the latch unit 20 and the input unit 30 toor from the high-potential power source Vcc according to the logics ofthe signals Axm1 and Axm2, and a switch 43 a for connecting ordisconnecting the input unit 30 to or from the ground power source Vssaccording to the logic of the signal A.

Note that, with the above configuration, the number of switches used forconnecting or disconnecting the high-potential power source Vcc to orfrom the source of the N-type transistor of which the gate is connectedto the signal Vi− has been two, and the number of switches used forconnecting or disconnecting the high-potential power source Vcc to orfrom the source of the N-type transistor of which the gate is connectedto the signal Vi+ has been two, but each of these is not restricted totwo, and rather may be multiple of two or more. In this case, the signalAxpn (n is a positive integer equal to or greater than 2), and thesignal Axmn (n is a positive integer equal to or greater than 2), to beconnected to each of the switches are each independent, and thetrailing-edge period of the logic of each signal is also setindependently.

Thus, the disconnecting periods of the switches 41 b, 41 c, 42 b, and 42c are adjusted by the comparative operation control circuit 640, wherebythe lowering speeds of the potentials of the signals Vm and Vp betweenthe latch unit 20 and the input unit 30 can be adjusted at the time ofcomparing the potentials of the signals Vi+ and Vi−. Thus, thedifference between the potential of the signal Vi+ and the potential ofthe signal Vi−, and correlation between the trailing-edge period of thesignal A, and the trailing-edge periods of the signals Axp1 and Axp2,shown in FIG. 13 can be adjusted. Note that it goes without saying thateven when there are multiple switches, multiple signals Axp, andmultiple signals Axm, the disconnecting period of each switch can beadjusted.

Thus, the ADC 600 according to the seventh embodiment is a successivecomparative analog-to-digital conversion device including a comparatorincluding an input unit (input unit 30) for receiving a complementaryinput signal to generate a inversion complementary signal having theinversion logic of the complementary input signal, and a latch unit(latch unit 20) for latching the inversion complementary signal; acomparative operation control circuit 640 including multiple firstswitches (switches 41 b and 41 c) for connecting or disconnecting thelatch unit and the input unit to or from the high-potential power sourceaccording to the logic of each of multiple first signals (signals Axp orsignals Axm), and multiple second switches (switches 42 b and 42 c) forconnecting or disconnecting the latch unit and the input unit to or fromthe high-potential power source according to the logic of each ofmultiple second signals (signals Axp or signals Axm); a delay circuit520 for outputting each of the first signals (signals Axp or signalsAxm), and each of the second signals (signals Axp or signals Axm); and asuccessive operation control circuit for outputting a control signalwhich controls the period of logic change of each of the first signalsfor controlling disconnection of each of the multiple first switches(switches 41 a), and the period of logic change of each of the secondsignals for controlling disconnection of each of the multiple secondswitches (switches 42 a).

Eighth Embodiment

With the sixth embodiment, the signals Axp and Axm are output from aninverter for corrugating disposed on the subsequent stage of an inverterfor generating signal delay at the delay circuit 520. On the other hand,with the eighth embodiment, the signals Axp and Axm are output from aninverter for generating signal delay at a delay circuit 720.

FIG. 18 illustrates an ADC circuit 700 according to the eighthembodiment. The ADC 600 according to the seventh embodiment includes alatch unit 20, an input unit 30, a comparative operation control circuit40, a delay circuit 720, a successive comparative operation controlcircuit 530, and a sample-hold circuit 540.

The latch unit 20, input unit 30, comparative operation control circuit40, delay circuit 720, successive comparative operation control circuit530, and sample-hold circuit 540 are the same circuits as described withthe sixth embodiment, and description regarding configuration andoperation thereof will be omitted.

The delay circuit 720 includes inverters 721, 722, 723, 724, 725, 726,and 727.

The inverter 727 receives the signal CLK and outputs the inversionsignal thereof. The inverters 723 and 726 receive the signal to beoutput from the inverter 727, and output the inversion signal thereof.The inverters 722 and 725 receive the output signals from the inverters723 and 726 and output the inversion signals thereof, respectively. Theinverters 721 and 724 receive the signals to be output from theinverters 722 and 725 and output the inversion signals thereof as thesignals Axp and Axm.

With the inverters 721 and 724, delay time from input of the inputsignal to output of the output signal varies according to the signalCNTL531.

Thus, with the delay circuit 720, time difference from the trailing edgeof the signal A to the trailing edge of the signal Axp, and timedifference from the trailing edge of the signal A to the trailing edgeof the signal Axm are the same as with the delay circuit 520. However,with the delay circuit 720, the period of potential change at the timeof the trailing edges of the logics of the signals Axp and Axm becomeslong. This is because the signals output from the inverters 721 and 725become the signals Axp and Axm as is without passing through an inverterfor corrugating, and accordingly, the potential changes of the signalsAxp and Axm become moderate.

Thus, time from start of disconnecting operation to end thereof betweenthe high-potential power source Vcc, and the latch unit 20 and the inputunit 30 by the switch 41 a of the comparative operation control circuit40 is longer than time required for disconnecting operation by theswitch 41 a at the ADC 500.

As a result thereof, the falling speeds of the potentials of the signalsVm and Vp between the latch unit 20 and the input unit 30 changes at thetime of comparing the potentials of the signals Vi+ and Vi− as comparedto the case of the ADC 500 according to the first embodiment.

Thus, the difference between the potential of the signal Vi+ and thepotential of the signal Vi−, and correlation between the trailing-edgeperiod of the signal A and the trailing-edge periods of the signals Axm1and Axm2 and the signals Axp1 and Axp2, shown in FIG. 13, can beadjusted.

Thus, the ADC 700 according to the eighth embodiment is a successivecomparative type analog-to-digital conversion device including acomparator including an input unit (input unit 30) for receiving acomplementary input signal to generate a inversion complementary signalhaving the inversion logic of the complementary input signal, and alatch unit (latch unit 20) for latching the inversion complementarysignal; a comparative operation control circuit 40 including a firstswitch (switch 41 a) for connecting or disconnecting the latch unit andthe input unit to or from a high-potential power source according to thelogic of a first signal (signal Axp or signal Axm), and a second switch(switch 42 a) for connecting or disconnecting the latch unit and theinput unit to or from the high-potential power source according to thelogic of a second signal (signal Axp or signal Axm); a delay circuit 520including a first inverter (inverter 721) for outputting the firstsignal (signal Axp or signal Axm), and a second inverter (inverter 724)for outputting the second signal (signal Axp or signal Axm); and asuccessive operation control circuit for outputting a control signal forcontrolling the period of logic change of the first signal forcontrolling disconnection of the first switch (switch 41 a), and theperiod of logic change of the second signal for controllingdisconnection of the second switch (switch 42 a) based on the logic of asignal latched by the latch unit.

The first inverter and the second inverter according to the eighthembodiment include a logic inversion circuit for inverting the logic ofan input signal, and a circuit for adding load capacitance to an outputsignal line of the logic inversion circuit according to the binarynumeral represented by the control signal.

The first inverter and the second inverter according to the eighthembodiment include a logic inversion circuit for inverting the logic ofan input signal, and a circuit for varying resistance between the logicinversion circuit and the ground power source line according to thebinary numeral represented by the control signal.

Ninth Embodiment

With the sixth embodiment, the signals Axp and Axm are output from aninverter for corrugating disposed on the subsequent stage of an inverterfor generating signal delay at the delay circuit 520. On the other hand,with the ninth embodiment, the signals Axp and Axm are output from aninverter for generating signal delay at a delay circuit 820.

FIG. 19 illustrates an ADC circuit 800 according to the ninthembodiment. The ADC 800 according to the ninth embodiment includes alatch unit 20, an input unit 30, a comparative operation control circuit40, a comparative operation control circuit 840, a delay circuit 820, asuccessive comparative operation control circuit 530, and a sample-holdcircuit 540.

The latch unit 20, input unit 30, successive comparative operationcontrol circuit 530, and sample-hold circuit 540 are the same circuitsas described with the sixth embodiment, and description regardingconfiguration and operation thereof will be omitted.

The delay circuit 820 includes inverters 821, 822, 823, 824, 825, 826,and 827. The inverter 827 receives the signal CLK and outputs theinversion signal thereof to the inverters 823 and 826. The inverter 827outputs the signal A, signal Axp1, and signal Axm1. The inverters 823and 826 receive the output signal from the inverter 827 and output theinversion signal thereof to the inverters 822 and 825. The inverters 822and 825 receive the output signals from the inverters 823 and 826 andoutput the inversion signal thereof to the inverters 821 and 824. Theinverter 821 outputs the signal Axp2 to the comparative operationcontrol circuit 840, and the inverter 824 outputs the signal Axm2 to thecomparative operation control circuit 840.

The comparative operation control circuit 840 includes switches 41 b and41 c for connecting or disconnecting the latch unit 20 and the inputunit 30 to or from the high-potential power source Vcc according to thelogics of the signals Axp1 and Axp2, switches 42 b and 42 c forconnecting or disconnecting the latch unit 20 and the input unit 30 toor from the high-potential power source Vcc according to the logics ofthe signals Axm1 and Axm2, and a switch 43 a for connecting ordisconnecting the input unit 30 to or from the ground power source Vssaccording to the logic of the signal A.

Thus, disconnection between the latch unit 20 and input unit 30, and thehigh-potential power source Vcc by the switches 41 b and 42 b, andconnection between the input unit 30 and the ground Vss by the switch 43a are simultaneously executed by the delay circuit 820. Also, afterdelay time according to the binary numeral represented by the signalCNTL531 of the successive comparative operation circuit 530 by the delaycircuit 820, disconnection between the latch unit 20 and input unit 30,and the high-potential power source Vcc by the switches 41 c and 42 c isexecuted.

Thus, the disconnection periods of the switches 41 c and 42 c areadjusted by the delay circuit 820, and thus, when comparing thepotentials of the signals Vi+ and Vi−, the falling speed of thepotentials of the signals Vm and Vp between the latch unit 20 and theinput unit 30 can be adjusted. Thus, the difference between thepotential of the signal Vi+ and the potential of the signal Vi−, andcorrelation between the trailing-edge period of the signal A, and thetrailing-edge periods of the signals Axm1 and Axm2 and the signals Axp1and Axp2, shown in FIG. 13 can be adjusted.

Thus, the ADC 800 according to the ninth embodiment is a successivecomparative analog-to-digital conversion device including a comparatorincluding an input unit (input unit 30) for receiving a complementaryinput signal to generate a inversion complementary signal having theinversion logic of the complementary input signal, and a latch unit(latch unit 20) for latching the inversion complementary signal; acomparative operation control circuit 840 including a first switch(switch 41 c) for connecting or disconnecting the latch unit and theinput unit to or from a high-potential power source according to thelogic of a first signal (signal Axp2 or signal Axm2), a second switch(switch 42 c) for connecting or disconnecting the latch unit and theinput unit to or from the high-potential power source according to thelogic of a second signal (signal Axp2 or signal Axm2), and a thirdswitch (switch 41 b) and a fourth switch (switch 42 b) for connecting ordisconnecting the latch unit and the input unit to or from thehigh-potential power source according to the logic of a third signal(signal Axp1 or signal Axm1); a delay circuit 820 for outputting thefirst signal (signal Axp2 or signal Axm2), the second signal (signalAxp2 or signal Axm2), and the third signal (signal Axp1 or signal Axm1);and a successive operation control circuit 530 for outputting a controlsignal for controlling the period of logic change of the first signalfor controlling disconnection of the first switch (switch 41 c), and theperiod of logic change of the second signal for controllingdisconnection of the second switch (switch 42 c) based on the logic of asignal latched by the latch unit.

Tenth Embodiment

With the sixth embodiment, the sample-hold circuit 540 samples and holdsthe potential of the signal Vi+ and the potential of the signal Vi−making up the input signal Vin. On the other hand, an ADC 900 accordingto the tenth embodiment includes no sample-hold circuit 540. The reasonthereof is because with the tenth embodiment, sample-hold of thepotential of the input signal Vi is executed at a circuit for inputtingthe input signal Vi to the ADC 900 illustrated by the tenth embodiment.

FIG. 20 is a diagram illustrating the ADC 900 according to the tenthembodiment. The ADC 900 includes a latch unit 20, an input unit 30, acomparative operation control circuit 40, a delay circuit 520, and asuccessive comparative operation control circuit 530. Accordingly, theADC 900 differs from the ADC 500 in that the sample-hold circuit 540 isnot included.

Thus, the ADC 900 includes no sample-hold circuit 540, whereby the areaoccupied with circuits can be reduced as compared to the ADC 500according to the sixth embodiment.

Thus, the ADC 900 according to the tenth embodiment is a successivecomparative type analog-to-digital conversion device including acomparator including an input unit (input unit 30) for receiving acomplementary input signal to generate a inversion complementary signalhaving the inversion logic of the complementary input signal, and alatch unit (latch unit 20) for latching the inversion complementarysignal; a comparative operation control circuit 40 including a firstswitch (switch 41 a) for connecting or disconnecting the latch unit andthe input unit to or from a high-potential power source according to thelogic of a first signal (signal Axp or signal Axm), and a second switch(switch 42 a) for connecting or disconnecting the latch unit and theinput unit to or from the high-potential power source according to thelogic of a second signal (signal Axp or signal Axm); a delay circuit 520for outputting the first signal (signal Axp or signal Axm) and thesecond signal (signal Axp or signal Axm); and a successive operationcontrol circuit for outputting a control signal for controlling theperiod of logic change of the first signal for controlling disconnectionof the first switch (switch 41 a), and the period of logic change of thesecond signal for controlling disconnection of the second switch (switch42 a) based on the logic of a signal latched by the latch unit.

Eleventh Embodiment

FIG. 21 is a diagram illustrating a signal processing device 1 (receiver1) using the ADC illustrated by the sixth embodiment through tenthembodiment. The signal processing device 1 is a device which includes anantenna 2, a filter circuit and amplifier 3, an ADC 4, and a DSPdemodulator 5, and outputs a signal for propagating audio data or imagedata before modulation which is available at a display device 6, audiogenerating device 7, or the like.

The signal processing device 1 is a device for restoring the modulatedsignal received by the antenna 2 into the original signal. The filtercircuit and amplifier 3 is a circuit for amplifying the modulated signalby attenuating noise thereof. The ADC circuit 4 is a circuit forconverting the input modulated signal into a digital signal. Note thatthe ADC circuit 4 is the ADC circuit according to the sixth throughtenth embodiments. The DSP demodulator 5 receives the signal digitizedby the ADC circuit 4, restores the signal before modulation, and outputsthis to the display device 6 or audio generating device 7. Here, thesignal before modulation means a signal relating to image data for thedisplay device 6, a signal relating to audio for the audio generatingdevice 7, or the like.

Thus, with the ADC 4, when executing successive comparative operation todetect the difference between the potential of the signal Vi+ and thepotential of the signal Vi− making up the input signal Vi, a comparativeoperation control circuit is used, whereby the number of elements makingup the circuits can be reduced. Accordingly, with the whole receiver 1,the circuit area of the ADC 4 can be reduced, and accordingly, thecircuit area of the whole receiver 1 can be reduced.

Thus, the signal processing device according to the eleventh embodimentis a receiver (receiver 1) including a filter circuit (filter circuitand amplifier 3) for attenuating noise from an analog reception signal;an amplifier (filter circuit and amplifier 3) for amplifying the analogreception signal of which the noise is attenuated; an analog-tot-digitalcircuit according to one of the sixth through tenth embodiments forconverting the analog reception signal of which the noise is attenuatedinto a digital signal; a DSP demodulator (DSP demodulator 5) forrestoring the signal before modulation from the reception signal ofwhich the noise is attenuated.

All examples and conditional language recited herein are intended forpedagogical purposes to aid the reader in understanding the inventionand the concepts contributed by the inventor to furthering the art, andare to be construed as being without limitation to such specificallyrecited examples and conditions, nor does the organization of suchexamples in the specification relate to a depicting of the superiorityand inferiority of the invention. Although the embodiments of thepresent invention have been described in detail, it should be understoodthat the various changes, substitutions, and alterations could be madehereto without departing from the spirit and scope of the invention.

1-17. (canceled)
 18. An analog-to-digital conversion device comprising:a comparator including an input circuit for receiving a complementaryinput signal to generate an inversion complementary signal having theinversion logic of the complementary input signal, and a latch circuitfor latching the inversion complementary signal; a comparative operationcontrol circuit including a first switch for connecting or disconnectingthe latch circuit and the input circuit to or from a high-potentialpower source according to the logic of a first signal, and a secondswitch for connecting or disconnecting the latch circuit and the inputcircuit to or from a high-potential power source according to the logicof a second signal; a delay circuit for outputting the first signal andthe second signal; a successive operation control circuit for outputtinga control signal for controlling the period of logic change of the firstsignal for controlling disconnection of the first switch, and the periodof logic change of the second signal for controlling disconnection ofthe second switch, based on the logic of a signal latched by the latchcircuit.
 19. The analog-to-digital conversion device according to claim18, further comprising: a circuit for sampling and holding one signaland the other signal of the complementary input signal.
 20. Theanalog-to-digital conversion device according to claim 18, wherein thedelay circuit comprises: a first circuit for outputting the firstsignal; a second circuit for outputting the second signal; wherein thefirst circuit and the second circuit include an inverter circuit; andwherein the inverter circuit includes a logic inversion circuit forinverting the logic of an input signal, and a circuit for adding loadcapacitance to an output signal line of the logic inversion circuitaccording to the binary numeral represented by the control signal. 21.The analog-to-digital conversion device according to claim 18, whereinthe delay circuit comprises: a first circuit for outputting the firstsignal; a second circuit for outputting the second signal; wherein thefirst circuit and the second circuit include an inverter circuit; andwherein the inverter circuit includes a logic inversion circuit forinverting the logic of an input signal, and a circuit for makingresistance between the logic inversion circuit and a ground power sourceline variable according to the binary numeral represented by the controlsignal.
 22. An analog-to-digital conversion device comprising: acomparator including an input circuit for receiving a complementaryinput signal to generate an inversion complementary signal having theinversion logic of the complementary input signal, and a latch circuitfor latching the inversion complementary signal; a comparative operationcontrol circuit including a plurality of first switches for connectingor disconnecting the latch circuit and the input circuit to or from ahigh-potential power source according to the logic of each of aplurality of first signals, and a plurality of second switches forconnecting or disconnecting the latch circuit and the input circuit toor from a high-potential power source according to the logic of each ofa plurality of second signals; a delay circuit for outputting each ofthe first signals and each of the second signals; a successive operationcontrol circuit for outputting a control signal for controlling theperiod of logic change of each of the first signals for controllingdisconnection of each of the plurality of first switches, and the periodof logic change of each of the second signals for controllingdisconnection of each of the plurality of second switches, based on thelogic of a signal latched by the latch circuit.
 23. An analog-to-digitalconversion device comprising: a comparator including an input circuitfor receiving a complementary input signal to generate an inversioncomplementary signal having the inversion logic of the complementaryinput signal, and a latch circuit for latching the inversioncomplementary signal; a comparative operation control circuit includinga first switch for connecting or disconnecting the latch circuit and theinput circuit to or from a high-potential power source according to thelogic of a first signal, and a second switch for connecting ordisconnecting the latch circuit and the input circuit to or from ahigh-potential power source according to the logic of a second signal; adelay circuit including a first inverter for outputting the firstsignal, and a second inverter for outputting the second signal; asuccessive operation control circuit for outputting a control signal forcontrolling the period of logic change of the first signal forcontrolling disconnection of the first switch, and the period of logicchange of the second signal for controlling disconnection of the secondswitch, based on the logic of a signal latched by the latch circuit. 24.The analog-to-digital conversion device according to claim 23, whereinthe first inverter and the second inverter comprises: a logic inversioncircuit for inverting the logic of an input signal; a circuit for addingload capacitance to an output signal line of the logic inversion circuitaccording to the binary numeral represented by the control signal. 25.The analog-to-digital conversion device according to claim 23, whereinthe first inverter and the second inverter comprises: a logic inversioncircuit for inverting the logic of an input signal; a circuit for makingresistance between the logic inversion circuit and a ground power sourceline variable according to the binary numeral represented by the controlsignal.
 26. An analog-to-digital conversion device comprising: acomparator including an input circuit for receiving a complementaryinput signal to generate an inversion complementary signal having theinversion logic of the complementary input signal, and a latch circuitfor latching the inversion complementary signal; a comparative operationcontrol circuit including a first switch for connecting or disconnectingthe latch circuit and the input circuit to or from a high-potentialpower source according to the logic of a first signal, a second switchfor connecting or disconnecting the latch circuit and the input circuitto or from a high-potential power source according to the logic of asecond signal, and third and fourth switches for connecting ordisconnecting the latch circuit and the input circuit to or from ahigh-potential power source according to the logic of a third signal; adelay circuit for outputting the first signal, the second signal, andthe third signal; a successive operation control circuit for outputtinga control signal for controlling the period of logic change of the firstsignal for controlling disconnection of the first switch, and the periodof logic change of the second signal for controlling disconnection ofthe second switch, based on the logic of a signal latched by the latchcircuit.
 27. A signal processing device comprising: a filter circuit forattenuating noise from an analog reception signal; an amplifier foramplifying the analog reception signal of which the noise is attenuated;a DSP demodulator for demodulating a signal before modulation from thereception signal of which the noise is attenuated; an analog-to-digitalcircuit for converting the analog reception signal of which the noise isattenuated into a digital signal; wherein An analog-to-digitalconversion device comprises: a comparator including an input circuit forreceiving a complementary input signal to generate an inversioncomplementary signal having the inversion logic of the complementaryinput signal, and a latch circuit for latching the inversioncomplementary signal; a comparative operation control circuit includinga first switch for connecting or disconnecting the latch circuit and theinput circuit to or from a high-potential power source according to thelogic of a first signal, and a second switch for connecting ordisconnecting the latch circuit and the input circuit to or from ahigh-potential power source according to the logic of a second signal; adelay circuit for outputting the first signal and the second signal; asuccessive operation control circuit for outputting a control signal forcontrolling the period of logic change of the first signal forcontrolling disconnection of the first switch, and the period of logicchange of the second signal for controlling disconnection of the secondswitch, based on the logic of a signal latched by the latch circuit.